Rapid current demand microprocessor supply circuit

ABSTRACT

Apparatus are disclosed for controlling the delivery of power to DC components such as computer components, microprocessors or the like. Designs of voltage regulation modules are presented which are appropriate for faster components, lower voltages, and higher currents. Embodiments are especially suited to applications which cause rapid changes in the conductance of the load, even in the sub-microsecond time domain as is common in computer applications and the like and in powering electronics equipment, especially a distributed system and especially a system wherein low voltage at high current is required. Embodiments and sub-elements provide energy storage for low voltage, high current electronic loads, an ability to supply current with rapid time variation, providing extremely low inductance connections, permitting components to be located relatively remotely from the powered electronic load.

CROSS-REFERENCES TO RELATED APPLICATIONS

[0001] This application is a continuation application of, and claims thebenefit and priority of, U.S. patent application Ser. No. 10/030,379,filed Jan. 2, 2002, which is the United States National Stage ofInternational Application No. PCT/US00/18086, published, filed Jun. 30,2000, which claims the benefit of and priority from: (a) U.S.Provisional Application No. 60/142,102 filed Jul. 2, 1999; (b) U.S.Provisional Application No. 60/144,342, filed Jul. 16, 1999; (c) PCTApplication Number PCT/US00/07779, the specification of which was filedon Mar. 23, 2000 and designating the United States of America, this PCTApplication being filed while the Original US Application was pending;this PCT Application having been published, and InternationalApplication No. PCT/US00/18086 related as a Continuation-in-Part of PCTApplication Number PCT/US00/07779; (d) U.S. application Ser. No.09/534,641 filed Mar. 23, 2000; International Application NumberPCT/US00/18086 related as a Continuation-in-Part of U.S. applicationSer. No. 09/534,641, U.S. application Ser. No. 09/534,641 now issued asU.S. Pat. No. 6,307,757, U.S. application Ser. No. 09/534,641 itselfclaiming priority to each of U.S. Provisional Application 60/125,768(filed March 23, 1999) and U.S. Provisional Application 60/133,252(filed May 8, 1999); and (e) U.S. application Ser. No. 09/584,412 filedMay 31, 2000 and now issued as U.S. Pat. No. 6,694,438, InternationalApplication Number PCT/US00/18086 related as a Continuation of U.S.application Ser. No. 09/584,412; each hereby incorporated by reference.

I. TECHNICAL FIELD

[0002] This invention is applicable for use in powering a wide varietyof circuitry that requires low voltage and high current. In addition itprovides capability to provide rapidly changing current. In particularit applies to microprocessors and similar circuitry especially wherethey are requiring less than 2 volts and are projected to require lessthan one volt.

[0003] Buck converter topologies are in current use for poweringmicroprocessors. For a 2.5 volt, 13 ampere requirement, a switchingfrequency of 300 kHz is becoming inadequate. To meet substantial stepload changes a large output capacitance is becoming required. Asmicroprocessor voltage requirements move downward toward 1.0 volt at 50amperes, the prior art topologies become even less suitable. With a dropin voltage (and an attendant drop in differential voltage tolerance) of2.5 times, and an increase of current of 4 times, a larger outputcapacitor is now needed to maintain the required step response. Itbecomes increasingly difficult or impossible, however, to locate such alarge capacitor close to the microprocessor connections. In addition,the cost of this approach increases with decreasing voltage. Onesolution to this problem has been to increase the frequency of thevoltage regulation module. When the frequency increases in such anarrangement, however, the non-resonant edges of this waveform causeproblems such as the commutation of FET output capacitance and preventincreasing the switching frequency above about a megahertz. Thissituation is rapidly becoming serious as microprocessors and other lowvoltage electronics are being developed which are increasingly difficultto provide suitable power for. The present invention permits theachievement of power for such needs. It permits higher frequencies andcan be configured to handle higher currents.

[0004] This situation is rapidly becoming serious as microprocessors andother low voltage electronics are being developed which are increasinglydifficult to provide suitable power for.

[0005] As mentioned, this invention specifically relates to poweringcomputer systems. Here, often switch-mode DC is created to power theinternal components of the system. It has particular applicability innew designs where microprocessors have high demands and power changes.Such can relate to the area of powering low voltage, high currentelectronics. As mentioned, though, the invention is applicable in thefield of computing, and much of the following description is presentedin that context. It should be understood, however, that otherembodiments are in no way limited to the field of computing, and areapplicable to a wide variety of circumstances wherein a variety of powerabsorbing loads which absorb electrical power may abruptly change theirpower absorbing characteristics (that is to say, their impedance mayundergo a rapid change). They are also applicable if such loads areseparated physically such that the voltage which may be dropped acrossthe dynamic impedance of the power carrying conductors is a significantfraction of the voltage delivered to such loads. They are alsoincreasingly applicable to applications wherein design tradeoffs areforcing a steady decrease in operating voltages. Such situations mayarise in telecommunications, radar systems, vehicle power systems andthe like, as well as in computing systems. Further, the DC/AC converteritself may have applications in broader and other contexts as well.

II. BACKGROUND

[0006] The architecture of computing systems has undergone tremendouschanges in the recent past, due principally to the advance ofmicrocomputers from the original four-bit chips running at hundreds ofkilohertz to the most modern 32 and 64 bit microprocessors running athundreds of megahertz. As the chip designers push to higher and higherspeeds, problems arise which relate to thermal issues. That is, as thespeed of a circuit is increased, the internal logic switches must eachdischarge its surrounding capacitance that much faster. Since the energystored in that capacitance is fixed (at a given voltage), as the speedis increased, that energy, which must be dissipated in the switches, isdumped into the switch that many more times per second. Since energy persecond is defined as power, the power lost in the switches thereforeincreases directly with frequency.

[0007] On the other hand, the energy stored in a capacitance increasesas the square of the voltage, so a capacitor charged to two volts willstore only 44% of the energy stored in that same capacitor charged tothree volts. For this reason, a microcomputer designed to operate at twovolts will, when run at the same speed, dissipate much less power thanthe same microprocessor operating a three volts. So there is a tendencyto lower the operating voltage of microprocessors.

[0008] Other considerations cause the microprocessor to exhibit a lowermaximum speed if operated at a lower voltage as compared to a higheroperating voltage. That is, if a circuit is operating at full speed, andthe voltage on that circuit is simply reduced, the circuit will notoperate properly, and the speed of the circuit (the “clock speed”) mayhave to be reduced. To maintain full speed capability and still operateat lower voltage, the circuit often must be redesigned to a smallerphysical size. Also, as the size of the circuitry is reduced, and layerthickness is also reduced, the operating voltage may need to be loweredto maintain adequate margin to avoid breakdown of insulating oxidelayers in the devices. For the past few years, these steps have definedthe course of microprocessor design. Key microprocessor designers,seeking the maximum speed for their products, have therefore expendedconsiderable effort trading off the following considerations:

[0009] higher speed chips are worth more money;

[0010] higher speed chips must dissipate more heat;

[0011] there are limitations to removal of that heat;

[0012] lower voltages reduce the heat generated at a given speed; and

[0013] smaller devices run faster at a given voltage.

[0014] Of course, there are many, many important trade-offconsiderations beyond these, but the above list gives the basic elementswhich relate to some aspects of the current invention. The result ofthese considerations has been for the microprocessor designers toproduce designs that operate at lower and lower voltages. Early designsoperated at five volts; this was reduced to 3.3. to 3.0, to 2.7, to 2.3,and at the time of writing the leading designs are operating at 2.0volts. Further reductions are in store, and it is expected that futuredesigns will be operated at 1.8, 1.5, 1.3, 1.0, and even below one volt,eventually perhaps as low as 0.4 volts.

[0015] Meanwhile, advances in heat removal are expected to permitprocessors to run at higher and higher heat dissipation levels. Earlychips dissipated perhaps a watt; current designs operate at the 30 wattlevel, and future heat removal designs may be able to dissipate as muchas 100 watts of power generated by the processor. Since the powerdissipated is proportional to the square of the operating voltage, evenas the ability to remove heat is improved, there remains a tendency torun at lower operating voltages.

[0016] All of this is driven by the fundamental consideration: higherspeed chips are worth more money. So the designers are driven toincrease the speed by any and all means at their disposal, and thisdrives the size of the chips smaller, the voltages lower, and the powerup. As the voltage drops the current increases for a given power,because power is voltage times current. If at the same time improvementsin heat removal permit higher powers, the current increases stillfurther. This means that the current is rising very rapidly. Early chipsdrew small fractions of an ampere of supply current to operate, currentdesigns use up to 15-50 amperes, and future designs may use as much as100 amperes or more.

[0017] As the speed of the processors increase, the dynamics of theirpower supply requirements also increase. A processor may be drawing verylittle current because it is idling, and then an event may occur (suchas the arrival of a piece of key data from a memory element or a signalfrom an outside event) which causes the processor to suddenly startrapid computation. This can produce an abrupt change in the currentdrawn by the processor, which has serious electrical consequences.

[0018] Inductance is the measure of energy storage in magnetic fields.All current-carrying conductors have associated with their current amagnetic field, which represents energy storage. It is well known byworkers in the art that the energy stored in a magnetic field is halfthe volume integral of the square of the magnetic field. Since the fieldis linearly related to the current in the conductor, it may be shownthat the energy stored by a current carrying conductor is proportionalto half the square of the current, and the constant of proportionalityis called the “inductance” of the conductor. The energy stored in thesystem is supplied by the source of electrical current, and for a givenpower source there is a limit to the rate at which energy can besupplied, which means that the stored energy must be built up over time.Thus the presence of an energy storage mechanism naturally slows down acircuit, as the energy must be produced and metered into the magneticfield at some rate before the current can build up.

[0019] The available voltage, the inductance, and the rate of change ofcurrent in a conductor are related by the following equation, well knownby those skilled in the art:

V=L*∂I/∂t,

[0020] where L is the inductance of the conductor, and ∂I/∂t is the rateof change of current in the conductor.

[0021] This equation states that the voltage required to produce a givencurrent change in a load on a power system increases as the time scaleof that change is reduced, and also increases as the inductance of anyconnection to that load is increased. As the speed of microprocessors isincreased, the time scale is reduced, and as the available voltage isreduced, this equation requires the inductance to be droppedproportionally.

[0022] Normally, in powering semiconductor devices one does not need toconsider the inductance of the connections to the device, but with modemelectronics, and especially with microprocessors, these considerationsforce a great deal of attention to be brought to lowering the inductanceof the connections. At the current state of the art, for example,microprocessors operate at about two volts, and can tolerate a voltagetransient on their supply lines of about 7%, or 140 millivolts. Thesesame microprocessors can require that their supply current change at arate of at least one-third or even nearly one ampere per nanosecond, or3*10⁸ or 10⁹ amperes/second, respectively. The above equation indicatesthat an inductance of about 140 picohenries (1.4*10⁻¹⁰ H) and ½nanohenry, (5*10⁻¹⁰ H) will drop a voltage of 140 millivolts at thesetwo rates of current rise. To put this number in perspective, theinductance of a wire one inch in length in free space is approximately20 nanohenries, or 20,000 picohenries. While the inductance of aconnection can be reduced by paralleling redundant connections, tocreate a connection with an inductance of 140 picohenries withconductors about a centimeter long would require some 20 parallelconductors, and for instance a connection with an inductance of ½nanohenry would require nearly 100 parallel conductors.

[0023] The foregoing discussion implies that the source of low voltagemust be physically close to the microprocessor, or more generally theactive portion of a particular component, which in turn implies that itbe physically small. While it might be suggested that capacitors mightbe used to supply energy during the delay interval required for thecurrent in the conductors to rise, the intrinsic inductance of theconnections to the capacitors currently severely limits this approach.So the system designer is faced with placing the source of power veryclose to the processor to ensure that the processor's power source isadequately stable under rapid changes in current draw. This requirementwill become increasingly severe as the voltages drop still further andthe currents increase, because the former reduces the allowabletransient size and the latter increases the potential rate of change ofcurrent. Both factors reduce the permissible inductance of theconnection. This can force the designer to use smaller capacitors whichhave low inductance connections, and because the smaller capacitorsstore less energy, this drives the power system to higher frequencies,which adds costs and lowers efficiency.

[0024] The foregoing remarks are not limited in computers to the actualcentral microprocessor. Other elements of a modern computer, such asmemory management circuits, graphic display devices, high speed inputoutput circuitry and other such ancillary circuitry have been increasedin speed nearly as rapidly as the central processing element, and thesame considerations apply.

[0025] Many modern electronics circuitry, including computers, arepowered by switchmode power conversion systems. Such a system convertsincoming power from the utility line to the voltages and currentsrequired by the electronic circuitry by operation of one or moreswitches. In low power business and consumer electronics, such asdesktop personal computers, the incoming power is supplied as analternating voltage, generally 115 volts in the United States, and 220volts in much of the rest of the world. The frequency of alternation iseither 50 or 60 Hertz, depending upon location. Such utility power mustbe converted to low voltage steady (direct) current, or dc, andregulated to a few percent in order to be useful as power for theelectronic circuits. The device which performs such conversion is calleda “power supply”. While it is possible to create a low voltage regulatedDC power source using simple transformers, rectifiers, and linearregulators, such units would be heavy, bulky and inefficient. In theseapplications it is desirable to reduce weight and size, and thisapproach is unsuitable for this reason alone. In addition, theinefficiency of linear regulators is also unacceptable. Efficiency isdefined as the ratio of output power to input power, and a lowefficiency implies that heat is being developed in the unit which mustbe transferred to the environment to keep the unit cool. The lower theefficiency the more heat must be transferred, and this is itself areason for finding an alternate approach.

[0026] For these reasons, virtually all modern electronics circuitry ispowered by switchmode conversion systems. These systems typicallyoperate as follows. The incoming utility power is first converted tounregulated direct current by a rectifier. The rectified DC is thenconverted to a higher frequency, typically hundreds of kilohertz, byelectronic switches. This higher frequency power is then transformed bya suitable transformer to the appropriate voltage level; thistransformer also provides isolation from the utility power, required forsafety reasons. The resulting isolated higher frequency power is thenrectified again, and filtered into steady direct current for use by theelectronics. Regulation of the output voltage is usually accomplished bycontrol of the conduction period of the electronic switches. Theresulting power conversion unit is smaller and lighter in weight thanearlier approaches because the size and weight of the transformer andoutput filter are reduced proportionally to the increase in frequencyover the basic utility power frequency. All of this is well known in theprior art.

[0027] In a complex electronic system, various voltages may be required.For example, in a computer system the peripherals (such as disk drives)may require +12 volts, some logic circuits may require +5 volts,input/output circuits may additionally require −12 volts, memoryinterface and general logic may require 3.3 volts, and the centralmicroprocessor may require 2 volts. Standards have developed so that thecentral power source (the device that is connected directly to theutility power) delivers ±12 and +5 volts, and the lower voltages arederived from the +5 supply line by additional circuitry, called voltageregulation modules, or VRMs, near to the circuits that require the lowervoltage. These additional circuits convert the +5 volt supply to highfrequency AC power again, modifying the voltage through control of theperiod of the AC power, and again re-rectifying to the lower voltage dc.

[0028] The resulting overall system is complex and not very efficient,in spite of the use of switchmode technology. In a typical 200 wattcomputer system, four watts are lost in the initial rectification of theutility line, eight watts in the electronic switches, 2.5 watts in thetransformer, 20 watts in the output rectification and filtering, andfour watts in the connections between the center power supply and theelectronics boards. Thus 38.5 watts are lost in the conversion processfor the higher voltage electronic loads. Substantial additional lossesmay be sustained in the low voltage conversion process. A typical 50watt voltage regulation module, which may convert +5 volts at 10 amperesto +2 volts at 25 amperes for the microprocessor, will itself havelosses of about one watt each in the AC conversion and transformer, andten watts in the final rectification and filtering. Other voltageregulation modules will have losses almost as great, resulting in lossesfor the entire system which may be one-third of the power used. Someparticularly inefficient approaches may demonstrate efficiencies as lowas 50%, requiring that the input power circuits utilize twice the powerrequired by the actual final circuitry, and requiring that twice theheat be dissipated in the electronics (which must be removed by a fan)as is theoretically required by the actual operating circuitry.

[0029] This system evolved over the years and is not optimum for manycurrent uses, but persists because of inertia of the industry andbecause of the perceived benefit of maintaining industry standards onvoltages and currents as generated by the central power unit.

[0030] An analysis of current trends in the microprocessor industryclearly indicates that the current system will not be adequate for thefuture. These trends show that the current draw of critical elementssuch as the core microprocessor has been steadily increasing and willcontinue to do so into the future. Meanwhile, the operating voltage hasbeen steadily decreasing, dropping with it the allowable tolerance ofthe supply voltage in absolute terms. Finally, the rate of change ofprocessor current—the current slew rate—is increasing very rapidly, withsubstantial additional increases forecast for the near future. All ofthese factors mitigate against the current technology and require a newapproach to be adopted in the future. It has been reliably estimatedthat the current powering and other technology will not last more thanone additional generation of microprocessors, and since designers arecurrently at work on the generation following the next, it can be saidthat these designers are in the process of developing a microprocessorwhich cannot be powered by currently available technology.

[0031] A further problem in the prior art is the use of square waveelectronic conversion techniques. Such technology, known as PWM, forPulse Width Modulation, produces switch voltage waveforms which havesteeply rising edges. These edges produce high frequency powercomponents which can be conducted or radiated to adjacent circuitry,interfering with their proper operation. These high frequency powercomponents may also be conducted or radiated to other electronicequipment such as radio or television receivers, also interfering withtheir proper operation. The presence of such components requires carefuldesign of the packaging of the power system to shield other circuitryfrom the high frequency power components, and the installation ofexpensive and complex filters to prevent conduction of these componentsout of the power supply package on its input and output leads. What isneeded then, is a power conversion system which enables small, highlyefficient voltage regulation modules to be placed close to the point ofpower use, which is fast overall, and which is itself efficient and atleast as low in cost as the prior art technology it replaces.

III. DISCLOSURE OF INVENTION

[0032] It is an object of the present invention, therefore, to provide ameans for storing energy with lower inductance connections than could beachieved with the prior art. It is a further object of the presentinvention to provide a source of energy at low voltage and high currentwhich does not need to be placed in very close proximity to theelectronic load. Similarly, it is yet an another object of the inventionto provide a source of low voltage which can sustain the voltage acrossthe powered load even in the presence of high rates of change of currentdraw

[0033] It is also an object of the present invention, to provide a meansof converting utility power to high frequency alternating power forefficient distribution at higher efficiency than can be achieved usingexisting techniques. It is also an object to provide a means ofconverting high frequency AC power to the low DC voltages and high DCcurrents required by current and future electronics at higher efficiencythan can be achieved using current techniques. It is another object ofthe present invention to maintain that efficiency over a wide range ofload conditions.

[0034] A further object of the present invention is to provide a sourceof high frequency power which is substantially smaller than that of theprior art. Similarly it is an object to provide a source of low voltageat high current which is substantially smaller than that of the priorart to permit such a source to be placed in very close proximity to theelectronic load.

[0035] It is also an object of the preset invention to provide closercontrol of the output voltage of a power source, even for extremelyshort time periods. That is to say, it is an object to ease the task ofthe powering or of providing a power source so that it does not needsuch wide bandwidth and has a small transient response to changes inload. Thus an object is to provide a system with better transientresponse to changes in load.

[0036] It is a further object of the invention to provide a powerconversion system which stores less energy than that required by theprior art.

[0037] It is additionally an object of the present invention to providea power conversion system which can be produced at lower cost than theprior art.

[0038] It is also an object to address problems associated with the useof square wave electronic conversion techniques. It is yet anotherobject of the invention to reduce possible interference between thepower system and the electronics being powered, as well as with otherdevices in the vicinity of the powered electronics, by reducing the rateof rise of currents and the rate of fall of voltages in the powersystem. Similarly, an object is to provide power using smoothly varyingwaveforms in the power conversion circuitry.

[0039] It yet a further object of one embodiment of the presentinvention provide to power with the aforesaid objects being satisfied,yet operate at either a constant frequency or, through otherembodiments, to accommodate variable frequencies as well.

[0040] Another fundamental aspect of the invention is the potential forthe affirmative use of the transformer leakage inductance. This can benecessary as the DC output voltage requirement is lowered.

[0041] Another benefit of this invention involves the very nature of apower source. By incorporating some or all these elements it can bepossible to provide power remotely. By making the output capacitanceconsist of only the bypass capacitors necessary on the microprocessorpins, the circuit feeding the microprocessor assembly can haveessentially an inductive output.

[0042] Several features will be disclosed which taken together orseparately can allow the power conversion frequency to be increased toprovide a low stored energy approach to meet the high di/dt requirementsfor next generation low voltage requirements. Thus, yet other objectsinclude providing a circuit and method for providing power toelectronics with low voltage, high current and high di/dt requirements,providing substantially higher power conversion frequencies, providing acircuit which allows a reasonable amount of transformer leakageinductance and switching device capacitances, providing a circuit ormethod whereby the synchronous rectifiers (SR's) always switch with zerovoltage across the device, allowing high frequency operation, providinga circuit or method whereby the control signal to the SR operates in anon-dissipative fashion, allowing HF operation, and providing a reducedsize of the output capacitance through HF operation.

[0043] Accordingly, in one embodiment the present invention is directedto a system of energy storage which can store more energy and be placedphysically farther from the powered electronics, through the reductionof magnetic fields surrounding the electrical connections and themagnetic energy stored therein, thereby creating a faster respondingstorage and powering medium. The reduction of the magnetic fields andthe resulting reduction of inductance permits electronics to operate athigher speed, and the increased energy storage permits the poweringsystem to operate at lower speed. This reduction in powering systemfrequency may permit lower costs than could be obtained using highfrequency power systems.

[0044] Similarly, the present invention in another embodiment isdirected to a system of power conversion which eliminates many of theelements of the prior art, by distributing high frequency AC power to apoint near the loads, and performing a single conversion from AC to DCat the point of power consumption. In particular, the present inventionaddresses this latter AC to DC conversion and solution of the problemsrelated to conversion of higher voltage AC power to very low DC voltageswith good regulation and transient response.

[0045] The elimination of many of the redundant elements in the priorart approach not only increases efficiency by eliminating a power losselement, but also reduces cost by elimination of the cost of theelements removed from the system. The reduction of frequency alsoincreases the efficiency of the powering system, because at higherfrequencies switching losses become increasingly important and may equalor exceed all other losses. The present invention accomplishes many ofthese objects by providing a low inductance connection for energystorage elements which is not limited in length through the mechanism ofreducing the volume of the magnetic field surrounding the conductorsintermediate to the energy storage element and the powered electronics.

[0046] In yet another embodiment, the present invention distributes highfrequency smoothly varying or even sinusoidal waveforms, which exhibitrelatively low rates of voltage change for a given frequency, and muchlower than alternative AC approaches, such as distribution of squarewave or trapezoidal waveforms. The distribution of sinusoidal ACvoltage, rather than DC voltages as is usually done in the prior art,not only simplifies the central power unit, but also greatly simplifiesthe voltage regulation modules, reducing cost and raising efficiency.This approach also results in greatly reduced interference between thepower unit and adjacent circuitry, and simplifies the design and reducesthe cost of the line filters used to avoid conducted interference alongthe utility power lines. Also, distribution of low DC voltages (e.g., 5volts) results in relatively higher losses in the distribution wires andconnectors when compared to the use of medium voltage alternatingdistribution levels (e.g., 30 volts rms), which nevertheless remain safeto touch.

IV. BRIEF DESCRIPTION OF DRAWINGS

[0047]FIG. 1-1 shows a conventional computer power delivery system ofthe prior art.

[0048]FIG. 1-2 is a more detailed depiction of a computer power deliverysystem of the prior art.

[0049]FIG. 1-3 indicates the parts of the computer power delivery systemof the prior art that may be eliminated by the present invention.

[0050]FIG. 1-4 shows a computer power delivery system according to oneembodiment of the present invention.

[0051]FIG. 1-5 indicates an embodiment of the power conversion elementof the present invention.

[0052]FIG. 1-6 depicts another embodiment of the power conversionelement of the present invention.

[0053]FIG. 1-7 shows details of a switch drive according to the presentinvention.

[0054]FIG. 1-8 shows a rectifier circuit of the present invention.

[0055]FIG. 1-9 shows a variation of output voltage with changes in thevalue of a capacitance in one embodiment.

[0056]FIGS. 1-10 and 1-11 show two variations of the voltage across aload resistance as a function of the load resistance.

[0057]FIG. 1-12 shows another embodiment with a two switch configurationand various general elements.

[0058]FIGS. 1-13 and 1-14 are plots of voltage waveforms at variouslocations for two different loads, high and low respectively.

[0059]FIG. 3-1 shows a traditional buck converter of the prior art.

[0060]FIG. 3-2 shows a waveform of the center point of the buckconverter shown in FIG. 3-1.

[0061]FIG. 3-3 shows an embodiment of a transformer and rectifierportion according to the present invention.

[0062]FIG. 3-4 shows the voltage waveforms as they may exist at variouslocations in the circuit shown in FIG. 3-3.

[0063]FIG. 3-5 shows one gate drive embodiment for the SR's according tothe present invention.

[0064]FIG. 3-6 shows a circuit for voltage control on the primary sidewith a single switching design.

[0065]FIG. 3-7 shows a family of drain to source voltages as a functionof the control input voltage across the FET.

[0066]FIG. 3-8 shows a circuit for voltage control on the primary sidewith a dual switching design.

[0067]FIGS. 3-9a, b, c & d shows various synchronous rectificationcircuits according to the invention.

[0068]FIG. 3-10 shows a bulk capacitor and a by pass capacitorarrangement as applied to a microprocessor system in the prior art.

[0069]FIG. 3-11 shows an overall preferred embodiment of the inventionusing a single switch control element.

[0070]FIG. 3-12 shows an overall preferred embodiment of the inventionusing a dual switch control element.

[0071]FIG. 3-13 shows an overall preferred embodiment of importantaspects of the aspect of the design.

[0072]FIG. 3-14 shows yet another preferred embodiment of a voltageregulation module design using a variable capacitor for primary sideregulation.

[0073]FIG. 3-15 is a Smith chart showing a range of VRM input impedancesvs load current percentage for one design of the present invention.

V. MODE(S) FOR CARRYING OUT THE INVENTION

[0074] As can be easily understood, the basic concepts of the presentinvention may be embodied in a variety of ways. These concepts involveboth processes or methods as well as devices to or which accomplishsuch. In addition, while some specific circuitry is disclosed, it shouldbe understood that these not only accomplish certain methods but alsocan be varied in a number of ways. Importantly, as to all of theforegoing, all of these facets should be understood to be encompassed bythis disclosure.

[0075] In the prior art, the central power supply provides severalstandard voltages for use by the electronics. Referring to FIG. 1-1,utility power (101), typically at 110 or 220 volt nominal AC poweralternating at 50 or 60 cycles, is converted by power supply (106) tostandard DC voltages, usually ±12 and +5 volts. These voltages arebrought out of the power supply on flying leads, which form a kind ofdistribution system (107), terminated in one or more connectors (108)These standard voltages are useful directly for powering most of theinput/output circuitry (140) and peripherals (144), such as a hard disc,floppy disc, and compact disc drives. As the technology of centralprocessing unit (CPU) chip (141) has advanced, as discussed above, theoperating voltage of such chips has steadily been reduced in the questfor higher and higher operating speeds. This increase in processor speedeventually required an increase in speed of the dynamic random accessmemory (DRAM) chips (143) used to hold instructions and data for theCPU, and as a result the operating voltage of these DRAM chips has alsobeen reduced. Also, not all of the logic required to manage theinput/output functions and particularly the flow of data to and from theCPU and the memory and external devices is present on the CPU chip.These management functions, along with housekeeping functions (such asclock generation), interrupt request handling, etc., can be dealt withby the “chip set”, shown in FIG. 1-1 as logic management circuits (145).These circuits also have steadily increased in speed and havecorrespondingly required lower operating voltages.

[0076] The standard voltages are thus too high to properly power CPU(141), memory (143), and management circuits (145). These may allrequire different voltages, as shown in FIG. 1-1, where the actualvoltages shown are representative only. These different voltages mayeach be created by an individual Voltage Regulation Module (112) (VRM),which may reduce the voltage supplied by the power supply (106) to thevoltage required by the powered circuitry.

[0077] From an overall point of view, the prior art process ofdelivering power to a circuit load such as CPU (141) involves all of thepower processing internal to power supply (106), distribution system(107) and connectors (108), and the power processing internal to VRMunit (112). This overall process is shown in FIG. 1-2. Central powersupply (106), also called the “silver box”, uses switchmode technology,with processing elements (102), (103), (104), and (105). The voltageregulation module (VRM) also uses switchmode technology. It should beunderstood that the discussion provided applies to both components. Thusthe various features discussed in one context should be understood aspotentially being applicable to the other. Focusing upon the silver boxdesign only for purposes of initial understanding, it can be understoodthat utility power (101) enters the silver box and is converted tounregulated DC power by rectifier unit, or AC/DC converter (102). Theresulting DCpower is then re-converted to alternating current power at ahigher frequency by inverter unit (103) (also called a DC/AC converter).The higher frequency AC is galvanically connected to and is at thevoltage level of utility power (101). Safety considerations requireisolation from utility power (101), and as the required output voltageis much lower than that of utility power (101), voltage reduction isalso needed. Both of these functions are accomplished by transformer(104). The resulting isolated, low voltage AC is then rectified todirect or multiply direct current power output(s) by rectifier andfilter unit (105), distributed to the circuitry loads by distributionwiring (107) and connectors (108). As mentioned before, specificstandard voltages ±12 and +5 volts must be converted to lower voltagesfor CPU (141), memory (143) and management logic (145), by VRM unit(112). The standard DC voltage from power supply unit (106) (usually +5volts) is converted to alternating power again by DC/AC converter (109),transformed to the lower voltage by transformer (110), and re-rectifiedto the proper low voltage by AC/DC unit (111).

[0078] As the voltage of the delivered power to the circuit load isdecreased, the current increases, and as the speed of CPU (141) isincreased, the power system must be able to handle larger and largerrates of change of current as well. As discussed above, this requiresthe source of power, which for CPU unit (141) (and other low voltagecircuits) is VRM (112), to be close to the circuit load. While for thenear term designs the rate of change of current can be handled bycapacitive energy storage, for future designs at still lower voltagesand higher currents VRM unit (112) must be made extraordinarily small sothat it can be placed close to its circuit load, and also must operateat a very high frequency so that large amounts of energy storage are notrequired. The requirement for low energy storage is rooted in the twofacts that there is no physical room for the larger storage elements andno tolerance for their higher intrinsic inductance. Thus a requirementemerges that the frequency of VRM (112) must be increased.

[0079] Further, a glance at FIG. 1-2 indicates at least two redundantelements which can be eliminated. The established policy of distributingdirect current power requires rectifier and filter (105), and the needfor dropping the voltage to lower levels requires re-conversion of theDC to alternating current power by inverter (109). One of these isclearly redundant.

[0080] This opens the possibility of reduction of cost by eliminatingelements (105) and (109), and choosing to distribute alternating currentpower instead of direct current power. Of course, the AC improvement mayalso be configured with existing, traditional DC leads as well in ahybrid system if desired. Returning to the improvement, however, asmentioned before, the frequency of inverter (109) has had to increaseand will continue to increase, which requires, in the reduced system,that the frequency of inverter (103) be increased to a level adequate toserve the future needs of the system. FIG. 1-3 indicates these redundantelements.

[0081] Another redundancy exists in principle, between transformers(104) and (110), but the desire to provide isolated power in thedistribution system (107) mandates the use of transformer (104), and therequirement for different voltages for the different loads may alsorequire the various VRMs to utilize transformer (110). Assuming thatthese elements are left in place, then, the use of high frequency ACdistribution produces a system as shown in FIG. 1-4. Thus one embodimentis directed specifically to the simplified VRM. Such an arrangement alsopermits electrically remote location of power element (e.g. at locationswhere the lead inductance would have otherwise have come into play usingthe prior techniques).

[0082] In FIG. 1-4, central power supply (147) converts utility power(101) to DCpower by AC/DC converter (146). This DCpower is thenconverted to high frequency sinusoidal power by DC/AC converter (113).The sinusoidal power (or perhaps “substantially” or “approximately”sinusoidal power, as may be produced by even a less than ideal inverteror the like) is distributed to the location of use of the power, wherehigh frequency VRMs (118) convert the sinusoidal power to low voltage,high current power for the circuit loads such as CPU (141), input/outputcircuits (140), logic management circuits (145), and memory(143). Inthis approach, a VRM is required not only for the aforementioned lowvoltage circuits, but also for peripherals (144), since the DC power(likely +12 volt) requirement for these units is not supplied by thecentral power supply (106). (Note, the central power supply (106) maysupply only sinusoidal high frequency AC power in this approach). HighFrequency Transformer (114) may thus provide galvanic isolation and maytransform the voltage from constant voltage Sinusoidal DC/AC Converter(113) to a level considered safe to touch.

[0083] It is possible to organize a distribution system which provides aconstant current to the totality of the loads, or alternatively toprovide a constant voltage to those loads. The architecture of computersystems and other complex electronic systems with loads which requiremultiple voltages is more suited to the latter approach. That is, it isdesirable that the magnitude of the distributed AC voltage be maintainedvery close to constant against any output load variation, even on amicrosecond time scale. Thus, it can accommodate a variable load, namelya load which alters at levels which would have caused variation in thepower supplied in arrangements of the prior art. It may also beimportant to keep the Total Harmonic Distortion (THD) of the distributedAC voltage low, to reduce Electro-Magnetic Interference (EMI). It shouldbe noted, however, that the present invention may be modified to providea constant current as well. That is, as those of ordinary skill in theart would readily understand, it is possible to modify the describedcircuit so that a constant current is delivered into a load which variesfrom nominal to a short circuit, for use in constant currentapplications.

[0084] Converter (113) may be designed to provide a constant outputvoltage with low THD, independent of load. Some of the embodimentspresented herein depend upon being supplied with a constant input DCvoltage from converter (146). It would of course also be possible tocreate this constant distribution voltage by feedback internal toconverter (113), as an alternative, which then would not require aconstant input voltage from converter (146). The latterapproach—creating constant voltage through feedback—requires that thefeedback system have very high bandwidth (high speed) in order tomaintain the output voltage very close to constant against any outputload variation, even on a nanosecond time scale. This feedback approachmay be difficult and expensive to achieve, and the present invention isdirected to accomplishing a constant voltage from converter (113) by theintrinsic operation of the circuit, without feedback. This can besignificant because it can satisfy the needs of a system which has rapidenergy demands such as a rapid current demand of at least about 0.2amperes per nanosecond, at least about 0.5 amperes per nanosecond, atleast about 1 ampere per nanosecond, at least about 3 amperes pernanosecond, at least about 10 amperes per nanosecond, and even at leastabout 30 amperes per nanosecond and beyond. It also can be significantbecause it can permit reaction to a change in conditions very quickly,such as within:

[0085] less than about a period of a “Nyquist frequency” (e.g. theNyquist rate, that is the maximum theoretical rate at which sampling ortransmission of an event can occur for a feedback-type of system),

[0086] less than about two and a half times a period of a Nyquistfrequency,

[0087] less than about five times a period of a Nyquist frequency,

[0088] less than about ten times a period of a Nyquist frequency,

[0089] less than about twice a period of said alternating power output,

[0090] less than about four times a period of said alternating poweroutput,

[0091] less than about 200 nanoseconds,

[0092] less than about 500 nanoseconds,

[0093] less than about 1000 nanoseconds, and

[0094] less than about 2000 nanoseconds.

[0095]FIG. 1-5 shows one embodiment of a constant voltage high frequencypower source to accomplish the function of converter (113). Here DCpowersource (119) is the circuit representation of the constant voltage fromconverter (146), and load (128) represents the constellation of loadsconnected to distribution system (115) (including the effects ofconnectors (18) and distribution system) (115). The voltage from source(119) is converted to a constant current by inductor (120) and eithershunted by switch (122) when the switch is ON, or permitted to flow intonetwork (148), comprising the elements in parallel with switch (122)when the switch is OFF. The network thus acts as a response network,that is, one which acts after the switch has transitioned. The voltageacross switch (122) is approximately zero when switch (122) is ON and isdependent upon the response of network (148) when switch (122) is OFF.This response waveform, or “switch voltage waveform” is transformed bynetwork (48) to form the voltage across load (128). It turns out to bepossible to choose the values of elements (123), (124), (125), (126),and (127) such that the switch voltage is zero at the commencement ofthe interval of time when switch (122) is ON, independent of the valueof the conductance of load (128), at least within a nominal range ofconductance for load (128). This may be accomplished in the followingway. If the conductance of load (128) is very small (light loading),little current will flow in inductance (127), and its value will notstrongly affect the waveform across switch (122). Then the values ofelements (123), (124), (125), and (126) may be chosen to cause thewaveform across switch (122) to be approximately zero, or to be adesired fixed value, at the moment when switch (122) begins to conduct.Clear descriptions for the methodology for accomplishing this maybefound in U.S. Pat. Nos. 3,919,656 and 5,187,580. Once this has beenaccomplished, the conductance of load (128) may be changed to themaximum nominal value, and the value of inductor (127) chosen to returnthe value of voltage across switch (122)at the commencement of its ONperiod to the value chosen in the first step. This algorithm will resultin a circuit for which the value of the switch voltage waveform at thecommencement of the ON period of switch (122) is nearly independent ofthe value of the conductance of load (128), within the defined nominalrange. It also results in a circuit for which the shape of the switchvoltage waveform varies minimally over the range of the conductance ofload (128). A significant function of the network formed by elements(123), (124), (125), (126), and (127) is to form a sinusoidal waveformacross load (128). Since this is a linear passive network, namely, anetwork with no active elements (including but not limited to steeringdiodes, diodes generally, other active elements, or the like) or anetwork without some type of feedback element (an element which senses acondition and then responds to that condition as a result of a delayeddecision-type of result), if the shape of the switch voltage waveformdoes not change in any substantial way, and especially if thefundamental frequency component of the switch voltage waveform (theFourier component of the waveform at the operating frequency) does notchange substantially, for this circuit the value of the sinusoidalvoltage across load (128) will not change substantially. Thus selectionof the values of elements (123), (124), (125), (126), and (127) in thismanner results in a stable, constant, high frequency, pure sinusoidalvoltage across load (128), independent of the value of the conductanceof load (128), thereby accomplishing the objective of providing aconstant voltage to the distribution system. It should be noted that theoperation of this network to produce a constant output voltage is veryfast; abrupt changes in the conductance of load (128) anywhere over itsentire nominal range may be corrected in a few cycles of operation. Thisis much faster than typical feedback approaches could make the samecorrection and serves to provide a fast acting network, namely one whichdoes not suffer the existing delay in a feedback type of system.

[0096] A unique element of the invention is its high efficiency over theentire load range from a nominal load to an open circuit or from anominal load to a short circuit. (As one skilled in the art shouldunderstand, one way to achieve one as opposed to the other simplyinvolves altering the AC distribution system by one-quarter wavelength.)This comes about largely as a result of the constant switch waveformdescribed above. Since the voltage waveform changes but little over theload range, switching losses in the circuit are not affected by loadvariations. It should also be noted that all of the benefits of thisinvention are obtained without changing the frequency of operation.Thus, high efficiencies such as at least about 80%, at least about 85%,at least about 90%, at least about 95%, at least about 98% and even atleast about 99% efficiency and beyond can be attained.

[0097] Such a circuit, which provides a constant voltage sinusoidaloutput across a load (or even in not strictly “across” the load, moregenerically “to which the load is responsive” thus encompassing bitdirect and indirect responsiveness) which can vary at high speed,utilizing a single or multiple switch and a simple circuit, operating atconstant frequency, while maintaining high efficiency over the entireload range, is a unique aspect of this invention in the field of powerconversion.

[0098] Another unique element of the invention is in the nature of themethod of driving switch (122). As has been pointed out previously,efficiency is important in these applications, and it is desirable notto waste energy anywhere, including the circuit used to drive switch(122). It is in the nature of high frequency switches such as FieldEffect Transistors (FETs) that they have a large input capacitance.Circuits which change the voltage on the gate terminal in a square-wavemanner must first charge that capacitance to a voltage well above thegate threshold voltage for switch (122), turning ON the FET, and in theprocess deposit energy into that capacitance. It must then dischargethat capacitance to a voltage well below the gate threshold voltage forswitch (122), in the process absorbing the energy stored in the gatecapacitance. The power lost in the process is the energy stored in thegate capacitance, multiplied by the frequency of operation, and this canbe a substantial number. In the present invention this loss is avoidedby affirmatively utilizing the gate capacitance of switch (122), thuscoordinating the circuitry to the gate or capacitance. That is, theenergy stored in the gate capacitance during the period switch (122) isON is, in the present invention, stored in another element of the systemduring the period switch (122) is OFF, and is thereby available on thenext cycle to return the gate above the threshold voltage for the nextON period. This may be accomplished by “resonating” the gate capacitance(or the effective capacitance of the system) with a series or parallelinductor. The entire system may be tuned to coordinate with thefrequency of the output and the output capacitance of the switch.Referring to FIG. 1-7, FET switch (122) is depicted as an internalswitch device (139) with an explicit gate capacitance (138), shownseparately. Gate drive circuit (121), according to the currentinvention, contains inductor (136) connected in series (or in parallelas shown in the dotted-line alternate connection) (137) which isselected such that the reactance of inductor (136) or (137) is equal tothe reactance of capacitor (138) at the frequency of operation. In thisway the energy in the gate system is transferred from gate capacitor(138) to inductor (136) (or its alternate) (137) and back again eachcycle, and only the inevitable losses in the inductor and gateresistance need to be regenerated for each cycle.

[0099] In such a system the gate voltage is substantially sinusoidal. Itwill be obvious to those skilled in the art that the duty cycle of thesystem (that is, the fraction of the total period that switch (122) isON) is determined by the fraction of the sinusoidal cycle which issubstantially above the threshold voltage of switch (122). It will alsobe obvious that, while the duty cycle of switch (122) may be controlledby the magnitude of the sinusoidal signal, such an approach placeslimits on the available range of duty cycle, and also may result inlonger than desirable commutation times (that is, the fraction of thetotal period during which the switch is transitioning from the ON to theOFF state), which may increase the losses of switch (122) and therebyreduce the efficiency of the system. For this reason, the drive waveformfor switch (122) may be divided in the present invention into an ACportion (149) and a DC portion (150), and variation in the duty cycle ofswitch (122) may be controlled by varying the relative magnitude of theAC and DC components of the drive waveform for switch (122).

[0100] An alternative approach to constant voltage, high frequency powergeneration is shown in FIG. 1-6. Here again, DCpower source (119) is thecircuit representation of the constant voltage from converter (146), andload (128) represents the constellation of loads connected todistribution system (115) (including the effects of connectors (108) anddistribution system) (115). Switch (122) is placed in series withinductor (129), across source (119). The voltage across inductor (129)is transformed by transformer (131) and placed across the networkcomprised of elements (132), (133), (134), and (135). This networkproduces the output voltage appearing across load (128), which againrepresents the constellation of loads connected to distribution system(115) (including the effects of connectors (108) and distributionsystem) (115). Provided the values of the circuit elements are properlychosen, this output voltage will be independent of the value of theconductance of load (128), within a nominal range of such conductance.To create this independence, it is sufficient to select the values ofthe elements such that, as one example, the reactance of inductor (129)in parallel with the magnetizing inductance of transformer (131) isequal to the reactance of capacitor (130) in parallel with the adjunctoutput capacitance of switch (122) at the frequency of operation, thereactance of inductor (132) in series with the leakage inductance oftransformer (131) is equal to the reactance of capacitor (133) at thefrequency of operation; and the reactance of inductor (134) is equal tothe reactance of capacitor (35) at the frequency of operation. Selectionof the values of the circuit elements in this manner will result in astable, constant, high frequency, pure sinusoidal voltage across load(128), independent of the value of the conductance of load (128),thereby accomplishing the objective of providing a constant voltage tothe distribution system.

[0101] The necessity for the parallel resonant circuit formed byinductor (134) and capacitor (135) is reduced if the minimum loadconductance is not too close to zero. That is, the network comprised ofelements (134) and (135) has the function of providing a minimum load tothe generator so that the output waveform remains sinusoidal when load(128) is removed or reduced to a very low value. Should the applicationto which the invention is applied not present a load variation down tolow values, or if the requirement for low THD not be present at lightloads, the network comprised of elements (134) and (135) may bedispensed with. Alternatively, the network comprised of elements (134)and (135) may be reduced to a single element, which may be either aninductor or a capacitor, if the highest efficiency is not demanded.

[0102] It is generally possible to also dispense with inductor (129) byaffirmatively utilizing the magnetization inductance of transformer(131). Similarly, it is generally possible to dispense with inductor(132) by affirmatively utilizing the leakage inductance of transformer(131). This may be accomplished through the modification of theconstruction of transformer (131) in the manner well known to thoseskilled in the art.

[0103] As before, to attain high efficiency it is important toaffirmatively utilize the gate capacitance of switch (122), and all theremarks made above in reference to FIG. 1-7 apply to the embodiment ofFIG. 1-6 as well.

[0104] As mentioned earlier, and referring to FIG. 1-4, converter 113,operating together with AC/DC element (146), is designed to provide aconstant high frequency AC output voltage with low THD, independent ofload. It is VRM (118) which must convert this high frequency AC powerfrom power unit (147) to low voltage high current DC power for use bythe powered circuitry (145), (141), and (143). FIG. 1-8 shows oneembodiment of the rectifier portion of one embodiment of a VRM toaccomplish this conversion in accordance with the present invention.Input AC power from power unit (147) may also be processed further toenhance its stability before the rectification process, and this furtherprocessing is not shown in FIG. 1-4. The result of this processing is astable regulated AC input (177) to the rectifier circuit (178) shown inthe dotted box in FIG. 1-8.

[0105] Rectifier circuit (178) is comprised of transformer (179), whichin practice will exhibit leakage inductance caused by imperfect couplingbetween its primary and secondary windings. This leakage inductance maybe represented in general as an inductance in series with the primary orsecondary of the transformer. In FIG. 1-8, it is represented by inductor(180), which therefore may not be an actual component in the circuit,but rather simply a circuit representation of part of the realtransformer (179), as built. It should be noted that, should the naturalleakage inductance of transformer (179) be smaller than desirable forany reason, additional inductance may be added in series with itssecondary (or primary) to increase the natural value, as will beunderstood by those skilled in the art. For the purposes of thisdisclosure, inductor (180) may be considered to be the total of thenatural leakage inductance of transformer (179) and any additionaldiscrete inductance which may have been added for any purpose.

[0106] Diodes (83)rectify the AC output of transformer (179), and filterinductors (184) and filter capacitor (185) create a steady DC output forconsumption by the microprocessor or other electronic load (186). Forsmall output voltages, the voltage drop across the diodes (183) is toolarge relative to the output voltage, resulting in loss of efficiency.As a result diodes (183) may be profitably replaced by field effecttransistor (FET) switches, which can be manufactured to have a muchlower voltage drop. In this case the FET devices require a drive signalto determine their conduction period; the circuitry to do this is notshown in FIG. 1-8.

[0107] A second problem which arises as the output voltage is dropped isthe intrinsic leakage inductance of transformer (179). This inductance,which, together with other circuit inductance, is represented asinductor (180), and acts as a series impedance which increases theoutput impedance of the overall circuit. That is, there is a naturalvoltage division between the reactance of inductor (180) and loadimpedance (136), which requires an increased input voltage incompensation, if the output voltage is to remain constant over changesin the resistance of load (186). This voltage division causes the outputvoltage to be a strong function of the resistance of load (186), whichis another way of saying that the output impedance of the circuit is notsmall compared to the load resistance (186).

[0108] The diodes (183) shown in FIG. 1-8 would ideally conduct wheneverthe voltage on their anodes was positive with respect to their cathodes,and would not conduct when the voltage was in the opposite polarity.This is what is called zero voltage switching, or ZVS, because theswitching point, or transition, from the conducting to the nonconductingstate occurs at zero voltage point in the waveform. Operating an FETdevice at ZVS is an advantage, because the losses are lowered, since thedevice does not have to discharge energy from its output capacitance, orthe energy stored in capacitors (182), which are in parallel with theswitches. As the output current through load (186) increases, the timingfor the switches to produce ZVS must change, and may complicate the FETdrive circuitry. In the description of the figures which follow, weshall nevertheless assume that the switches are operated at ZVSconditions, or that a true diode is used.

[0109]FIG. 1-9 shows how the output voltage varies with changes in thevalue of capacitance (182) placed across diodes (183). These curves wereplotted for an operating frequency of 3.39 MHz. As may be seen in FIG.1-9, as the value of capacitances (182) are increased, the outputvoltage (that is, the voltage across load resistance) (186) first beginsto increase, but as the value of the capacitance (182) is increasedstill further, the voltage across load resistance (186) begins to dropagain. Thus there is an optimum value for the capacitances (182) whichobtains the highest voltage transfer function. In FIG. 1-9 two curvesare shown, curve (187) a value of inductance (180) of 40 nH, and curve(188) for a value of inductance (180) of 20 nH. Curve (187) shows that apeak in output voltage occurs at a value for capacitances (182) of about27 nF, while curve (188) shows that a peak occurs at a value forcapacitances (182) of about 86 nF. Note that these are not a factor oftwo apart (86/27>3) as would be the case if the values of capacitances(182) and inductor (180) satisfied the resonance condition since the twocurves are for values of inductor (180) which are a factor of two apart.This means that the condition for maximum output is not the same as forresonance at the frequency of the input power from generator (177). Thetwo capacitors (180) may be replaced by a single capacitor (181) in aparallel position across the secondary winding of transformer (179) andinductor (180), with the same result, although the current in the diodes(183) will not be the same in this case.

[0110]FIGS. 1-10 and 1-11 show the voltage across load resistance (186)as a function of the load resistance (186). The slope of these curves isa measure of the output impedance of the circuit (178). That is, if theslope is zero, the output impedance is zero, and the circuit exhibits“natural regulation” without feedback. Curve (189) in FIG. 1-10 andcurve (192) in FIG. 1-11 show that, for a value for capacitances (182)equal to the value which results in a peak in voltage across loadresistance (186), a slope of nearly zero is obtained, without feedback.That is, for a proper selection of the value of capacitances (182) inrelation to inductance (180), the voltage across load resistor (186)becomes relatively independent of the actual value of the load resistor(186)—the output is “naturally regulated”. It will be seen that theadvantage of “natural regulation”—regulation without feedback—is thatone does not need to wait for a feedback system to recognize a change inoutput voltage compared to a reference, and to change some parameterinternal to the circuit. Under the described conditions, the outputvoltage is held constant and maintained so within a cycle or two of theoperating frequency, which is short compared to stable feedback systems.

[0111] Thus a system has been described which produces a stable outputvoltage over a wide range of load resistances without feedback, evenunder conditions of rapid change in the load resistance. For systemswhich can tolerate the change in output shown in the figures, nofeedback is required. For systems which require tighter control of theoutput voltage under conditions of changing load, feedback may be added,and it will be noted that the teachings of the present invention reducethe requirement for action on the part of the feedback system,permitting simpler, faster, and less costly feedback circuits to beused.

[0112] As mentioned earlier, the circuit can be embodied in a variety ofmanners to achieve the overall goals of this invention. For example,referring to FIG. 1-12 as but one other example of a circuit design, ingeneral, the circuit can be understood. It may have any combination of avariety more generically stated elements. First, it may have a constantoutput element, such as the constant output voltage element (161). Inthis arrangement, the constant output element serves to maintain someoutput parameter as a constant regardless of a variation such as mayoccur from the variable load. As one skilled in the art would readilyunderstand, the parameter maintained may be selected from a greatvariety of parameters, including but not limited to parameters such as:

[0113] a substantially constant switch voltage output which issubstantially constant over all levels at which said variable loadexists practically,

[0114] a substantially constant load voltage input which issubstantially constant over all levels at which said variable loadexists practically,

[0115] a substantially constant switch voltage Fourier transform whichis substantially constant over all levels at which said variable loadexists practically,

[0116] a substantially constant switch voltage output waveform which issubstantially constant over all levels at which said variable loadexists practically,

[0117] a substantially constant switch voltage transition endpoint whichis substantially constant over all levels at which said variable loadexists practically, and

[0118] all permutations and combinations of each of the above

[0119] In the configuration shown, this constant output voltage element(161) has inductor L1 and capacitor C5 which may be tuned for seriesresonance at the fundamental frequency of operation, inductor L2 andcapacitor C6 which may be tuned for parallel resonance at thefundamental frequency of operation, and capacitors C7 & C8 arranged toform a half supply with low AC impedance as is common for a half bridgeconfiguration, with R5 representing the load to be powered. Of course,from these general principles, as a person of ordinary skill in the artwould readily understand, other designs can be configured to achievethis basic goal.

[0120] Second, the system can include a constant trajectory element suchas the constant trajectory element (162). In this arrangement, theconstant trajectory element serves to maintain the response waveform (oreven the Fourier component of the waveform) as substantially a constantregardless of a variation such as may occur from the variable load. Inthe configuration shown, this constant trajectory element (162) hasinductor L4 connected to a half supply (shown as capacitors C7 & C8). Itprovides a constant current at the time of transition from switch T1conducting or switch T2 conducting (or visa versa) where diode D2 andcapacitor C2 are adjunct elements of switch T1, and diode D3 andcapacitor C4 are adjunct elements of switch T2. The trajectory which ismaintained may even be held to one which present a continuous secondderivative of voltage with respect to time. As shown herein, designs mayalso be configured to achieve a constant end point. The end point may ormay not be zero, for instance, it may be desirable in certain designs tohave a non-zero end point. That type of a design may include values suchas: zero volts, a voltage which is less than a diode turn-on level, lessthan about 5% of said switch DC supply voltage, less than about 10% ofsaid switch DC supply voltage, less than about 20% of said switch DCsupply voltage, and less than about 50% of said switch DC supplyvoltage, each over all levels at which said variable load existspractically. Regardless, a constant result (trajectory, end point, orotherwise) can be important since it is the voltage at the moment ofswitch turn-on or the avoidance of turning on the body diode which canbe highly important. Again, from all these general principles, as aperson of ordinary skill in the art would readily understand, otherdesigns can be configured to achieve each of these basic goals. Designsmay thus provide a network which is substantially load independent andwhich provides a substantially trajectory fixed response. Further, anynonlinear transfer characteristics of any component, such as thevaractor capacitance nature of many switches, the nonlinear transfercharacteristics of a transformer, or the like, can be affirmativelyutilized by the network as well for an optimal result.

[0121] Third, the circuit may include an energy maintenance element,such as the energy maintenance circuit (163). In this feature, theenergy maintenance circuit (163) serves to maintain the energy needed asa constant regardless of a variation such as may occur from the variableload. In the configuration shown, this energy maintenance circuit (163)has a capacitor C6 configured in parallel with inductor L2, both beingin parallel with the load shown as R5. This element may serve to supplysubstantially all of the rapid energy demand of the load such asdiscussed earlier. Again, as before other designs can be configured toachieve this basic goal.

[0122] Fourth, the circuit may have some type of stabilizer element suchas the stabilizer element (164) shown. This stabilizer element(164)serves to absorb energy not in the fundamental frequency inaccordance with the principles discussed in U.S. Pat. No. 5,747,935,hereby incorporated by reference, to the assignee of the presentinvention.

[0123] Finally, the circuit may include an automatic bias network suchas the direct bias alteration element (165) as shown for each switch. Inthis arrangement, these networks may include some type of voltagedivider (166) with a conduction control element such as diode (167).Here, the voltage divider (166) uses two resistors R1 & R2 which may beselected to be equal, each with high values such as 1k ohm. This elementprovides a negative bias in proportion to the AC drive amplitude. Theresult can be a conduction period which is independent of the driveamplitude. It can thus provide a constant dead time (response time) whenneither switch is in the conductive state. Again, from these generalprinciples, as a person of ordinary skill in the art would readilyunderstand, other designs can be configured to achieve this basic goalas well.

[0124] As illustrated in FIGS. 1-13 and 1-14, it can be seen how aproperly configured system according to the present invention has theconstancy features mentioned. The plots 1-4 show waveforms as follows:

[0125] 1—the voltage at the junction between switches T1 and T2;

[0126] 2—the output voltage across the load, R5;

[0127] 3—the current through L1; and

[0128] 4—the current through L4.

[0129] By comparing the high load and low load situations for the samenetwork as shown between the two figures, several events can be noticed.These include the constant output voltage (A), constant end point (B andB′), constant trajectory (C and C′), constant response time period (Dand D′), zero voltage switching (B and B′), and constantly an event ofzero load current in the transition (E), all even though there is ahighly varying power and load current as indicated by the current intothe network at L1 (F and F′). Other features are also noticeable, as oneskilled in the art should easily understand.

[0130] As mentioned earlier, buck converter topologies (such as shown inFIG. 3-1) are in current use for powering microprocessors, especiallyfor voltage regulation modules. For a 2.5 volt, 13 ampere requirement, aswitching frequency of 300 kHz is becoming inadequate. To meetsubstantial step load changes an output capacitance (301) of 3 mF(millifarads) is becoming required. As microprocessor voltagerequirements move downward toward 1.0 volt at 50 amperes, the prior arttopologies become even less suitable. With a drop in voltage (and anattendant drop in differential voltage tolerance) of 2.5 times, and anincrease of current of 4 times, an output capacitor of 30 mF would beneeded to maintain the required step response. It becomes increasinglydifficult or impossible, however, to locate such a large capacitor closeto the microprocessor connections. In addition, the cost of thisapproach increases with decreasing voltage. The other possibility wouldbe to increase the frequency. The voltage waveform (302) shown in FIG.3-2 is typical for a buck converter. When the frequency increases insuch an arrangement, however, the non-resonant edges of this waveformcause problems such as the commutation of FET output capacitance andprevent increasing the switching frequency above about a megahertz. Thissituation is rapidly becoming serious as microprocessors and other lowvoltage electronics are being developed which are increasingly difficultto provide suitable power for. The present invention permits theachievement of higher frequencies and currents as will be required. Itpermits frequencies such as greater than at least about 300 kHz, greaterthan at least about 500 kHz, greater than at least about 1 MHZ, greaterthan at least about 3 MHZ, greater than at least about 10 MHZ, and evengreater than at least about 30 MHZ and beyond, and can be configured tohandle currents of more than about 15 amperes, more than about 20amperes, more than about 50 amperes, and even more than about 100amperes and beyond.

[0131] In one embodiment, an aspect of this invention is the basicchange from a circuit converting DC to DC to a circuit transforming ACto DC making use of a transformer and a synchronous rectifier. Atransformer is useful in this approach as it is possible to eliminatelarge currents being distributed to the converter input. The highcurrent secondary can thus be located physically close to the load. Onecircuit for accomplishing this shown in FIG. 3-3.

[0132] With the invention disclosed, the energy conversion frequency canbe increased substantially, thereby allowing the output capacitance(303) to remain small and be located adjacent to a given load such asthe microprocessor interconnections. In fact, much higher conversionfrequencies can be achieved and whereby the output capacitance can besubstantially reduced. In the case of the 1.0 volt, 50 ampererequirement, the output capacitance (303) with the present invention canbe 500 μF or lower, depending upon load requirements. In fact, with thepresent invention, designs can be accomplished which provide a networkhaving an effective capacitance (that which causes an appreciable effectin the use or circuit designed) which is less than about 10 millifarads,less than about 3 millifarads, less than about 1 millifarads, less thanabout 0.5 millifarads, and even less than about 0.3 millifarads.

[0133] Such a dramatic improvement can come through the incorporation ofseveral elements individually or simultaneously. One primary goal ofthis invention is the elimination of frequency related limitations.Consequently it can be important to eliminate forced voltage commutationof any capacitors. The Synchronous Rectifier (SR) (304) device used maybe a Field Effect Transistor (FET) with adjunct drain to sourcecapacitance (305). This SR can always be commutated to the conductingstate at a time when there is zero voltage across it.

[0134]FIG. 3-3 shows a preferred embodiment for the rectificationportion of a low voltage high current supply. The element LT (306)(total series inductance) is defined as the total of the transformerleakage inductance plus any other inductance in series with thetransformer (inductance in the primary is simply scaled to thesecondary). The element CT (total parallel capacitance) is defined asthe total of the SR adjunct capacitance (305) (Coss), plus any externalparallel capacitance of each SR (307) (Csr) plus any capacitance inparallel with the transformer secondary (308) (Cp).

[0135] There are several parameters which may be considered to optimizethis circuit. If the load being powered has the possibility of highdi/dt or if the load current can be a step function up or down then thefollowing parameters could be considered:

[0136] fundamental frequency of operation

[0137] transformer turns ratio

[0138] LT

[0139] CT

[0140] conduction angle (CA) for the SR's

[0141] phase delay (PD) of the SR's

[0142] The output inductance LF and capacitance CF can be important butmay have a less direct impact on the proper operation of the invention.

[0143] Also to be potentially considered is the basic relationshipbetween conduction angle and efficiency. In prior art and practice theconduction angle for the SR's has been carefully chosen to be less thanor equal to 180 degrees (i.e., no SR conduction overlap) to prevent ashort circuit on the transformer secondary. This common misperceptionarises from lower frequency assumptions. With the present invention, aconduction angle greater than 180 degrees is not only allowed butprovides a fundamental benefit of operation. Conduction angles in therange of 300 degrees or higher are clearly demonstrated. With properlychosen LT, CT, phase angle (PA) and conduction angle (CA), the drainwaveforms on the SR's (304) shown in FIG. 3-4 can be realized. Withthese conditions, a low ratio of SR root-mean-square (RMS) current tooutput current can be realized. Ratios of <1.3:1 have been achieved.

[0144] Just as a general comparison, the waveforms from FIG. 3-4 can becompared to FIG. 3-2 from the prior art. They both share the low dutycycle aspect but it is clear in FIG. 3-4 the switching of the SR occursat zero volts and is ideally lossless.

[0145] Leakage Inductance & Overlapping Conduction Angle:

[0146] The transformer leakage inductance is a fundamental limitingfactor for low voltage, high current, high frequency power supplies. Itconsists of an inductance in series with the transformer and hashistorically limited the conversion frequency.

[0147] In other art leakage inductance has been dealt with in variousways. Three patents, by Schlecht, Lee and Bowman, covering DC to DCconverters will be touched on as all include methods of handling theleakage inductance. In Schlecht et al., U.S. Pat. No. 4,788,634, theleakage inductance is managed by minimizing it. As that patent states:“It is desirable to limit the size of this leakage inductance to anegligibly small value compared to the resonant inductor (in this casethe transformer primary inductance) such that the unilateral conductingelement and controllable switch both have zero voltage switchingtransitions.” In Lee et al., U.S. Pat. No. 4,785,387 and Bowman U.S.Pat. No. 4,605,999 the transformer leakage inductance is used in acircuit resonant at or slightly above the fundamental frequency. Thegoal for this circuit is to accomplish zero voltage switching both forthe primary switches as well as for the rectifiers. However, the presentinvention shows use of the leakage inductance in a manner not resonantat the fundamental frequency.

[0148] One fundamental aspect of this invention is a circuit topologyand class of operation which can make allowance for a larger leakageinductance. This benefit can be realized by the choice of a highconduction angle in the SR's. In fact, for some applications conductionangles even greater than 300 degrees are shown to be valuable. As theoutput voltage requirement is reduced and the current requirement isincreased, both of these shifts result in still higher conductionangles. The setting of this large conduction angle, the total inductanceand total capacitance is done simultaneously with one of the desirableconditions being Zero Voltage Switching (ZVS) for the synchronousrectifiers. This allows operation at a higher frequency or, at a givenfrequency operation with a higher leakage inductance. This combinationof high frequency operation and/or higher leakage inductance toleranceis a fundamental benefit of this design and may perhaps be a necessarybenefit as microprocessor power requirements become more difficult tofulfill.

[0149] One additional note with respect to the total capacitance—thechoice of location between putting the capacitor across the transformer(308) or across the SR's (307) changes the current waveform through theSR's but does not greatly affect the voltage waveform. With thecapacitor across the transformer makes the current waveform more like asquare wave while it is quasi-sinusoidal when the capacitor is acrossthe SR. This difference can have significant ramifications as those ofordinary skill in the art should readily understand to some degree.

[0150] High Voltage on SR:

[0151] One general principal observed in rectifier circuit design is tominimize the reverse voltage stress across the rectifier device.Depending on the type of filter input the peak inverse voltage isusually in the range of being equal to the DC output voltage upwards to1.4 times the output voltage or in rare circumstances up to twice the DCoutput voltage.

[0152] One consequence though of the high conduction angle issubstantially higher voltage across the rectifier devices. For examplein the circuit values disclosed here the output voltage is 1.8 voltswhile the voltage across the rectifier devices is 15 volts! Historicallythis type of circuit performance has been thought of as poor practicefor a variety of reasons as those of ordinary skill in the art wellunderstand. Perhaps this is one reason why such a valuable circuit hasnot been discovered to date.

[0153] But a high conductance angle with attendant high voltage acrossthe SR during the non-conducting state has the benefit of low RMScurrent through the SR during the conducting state and is a conditionfor allowing large transformer leakage inductance. This circuit isideally suited for low voltage, high current requirements. Furthermoreit is well suited to loads which have a high di/dt requirement as aresult of the higher operating frequency and lower stored energy in theoutput capacitance. As it turns out the higher voltage requirement forthe SR's is not troublesome. With current manufacturing technology thereappears to little benefit to restraining the SR off state voltage toless than about 20 volts.

[0154] Gate Drive:

[0155] The next circuit being disclosed, FIG. 3-5, is a gate drivecircuit that derives its power from the AC input and uses only passiveelements. The gate drive of the SR's is also almost lossless. This allresults in low cost and predictable performance. It is also importantfor higher frequency operation.

[0156] In addition it is possible to add a DC or low frequency bias toprovide regulation or improve efficiency under various load conditions.In FIG. 3-5 the point labeled BIAS INPUT is an example of an injectionpoint for the control input. Varying the voltage on this input has theeffect of varying the conduction angle of the SR's without effecting theDELAY ANGLE (FIG. 3-4).

[0157] The correct phase angle for conduction of the SR's is determinedby the gate drive. Referring to FIG. 3-4, the angle labeled DELAY ANGLEcould be derived by using something like elements L1, R1,2 and C1,2 ofFIG. 3-5. The inductance L1 includes the gate drive transformer leakageinductance.

[0158] There could be many variations of gate drive which embody theseprinciples. This may be contrasted with conventional technology in whichthe gate drive is derived from a DC source and involves timing circuitryand switching devices.

[0159] Regulation with the SR:

[0160] It is possible to also control and/or regulate the output voltageby varying the SR Conduction Angle (CA). Consider FIG. 3-3 again withthe inclusion of the capacitor Cin 309 shown in dotted lines.

[0161] To select values for the controlled output circuit, first examinethe case where the CA goes to 360 degrees for the SR's. This results ina zero DC output. The impedance of Cin 309 should now be matched to thevalue of LT (transformed to the primary by the square of the turnsratio) forming a parallel resonant circuit at the fundamental frequency.As can now be seen the AC input is only loaded by a parallel resonantcircuit which in the ideal condition is lossless.

[0162] There exists a continuum of CA's from 360 degrees downward untilthe full load condition is reached as before. With properly chosencircuit parameters ZVS switching can be maintained over the wholeregulation range. One important requirement for ZVS is to provideconstant phase relationship between conduction time and the AC input. Inthe first order analysis, the only control input required is that shownin FIG. 3-5.

[0163] Parametric Regulation:

[0164] Another method of providing regulation or control of the outputcould be to use parametric elements such as a varactor capacitor orsaturable inductor to vary the output voltage. This can involve tuningthe circuit to maximize the sensitivity to a given element andsubsequently varying it. Another approach to this type of design is tobegin with a basic transfer function having the characteristic of avoltage source. Then with small changes in one or more variableelements, the output can be held constant.

[0165] For some load requirements, this method of control may be thesimplest or most cost effective. In particular loads which do not havehigh di/dt requirements or if the voltage required is not too low,parametric regulation may be ideal.

[0166] This method of control may have the disadvantage of poor responsetime for varying loads and poor input regulation. Another disadvantageis the incumbent increased sensitivity to component tolerances. In FIG.3-4 it can be seen that the CA is quite large. In general, the optimumCA increases for lower output voltages. One consequence when usingparametric regulation is that it can become increasingly difficult tomanage the increased sensitivity of the output voltage to the actualcircuit values. If the component sensitivity becomes unmanageable, itmay be preferable to optimize the rectification portion of the circuitfor rectification only and regulate or control on the primary side ofthe transformer, where the impedance is higher. Layout and componentvalues can be more manageable on the primary. Naturally linearcomponents such as linearly variable capacitors, linearly variableinductors, or even linearly variable resistors (as should be understood,resistors are likely not the preferred component since they may causelosses) may be utilized as well.

[0167] Regulation on the Primary Side (with a Single Ended Switch):

[0168]FIG. 3-6 shows a simplified series switch on the primary side ofthe transformer. This circuit design can be used to vary the AC voltageon the input of the transformer as a potential method of regulating theDC output. For instance, C1 (310) can be resonant with any residualinductive component of the rectifier circuit. C2 (311) may be lowimpedance at the fundamental frequency. The duty cycle of Q1 (312) canbe controlled to vary the AC voltage into the rectifier circuit. Thephase delay (313) (L1, R1, and C4) may be chosen such that at thecommencement of conduction the voltage across Q1 (312) is substantiallyzero. Further, the gate drive of Q1 can be set in similar fashion to thegate drive for the synchronous rectifier discussed earlier. The AC input(315) may be used as the source power, transformed down in voltage andsupplied to the gate through the delay circuit (313). In series withthis drive signal can be a control input (314). By summing these twovoltages the conduction angle can be varied from 0 to 360 degrees.

[0169] The conduction angle can be set by the control input and thephase relationship maybe derived from the AC input (315). With properlychosen circuit elements and delay time, Q1 (312) may be alwayscommutated to the conducting state at a time when the voltage across itis zero. Thus the AC voltage to the rectifier circuit can be varied fromnearly zero to full while maintaining a lossless condition. FIG. 3-7shows a family of voltage waveforms across Q1 (312) (Vds for a FETswitch) as a function of the control input. The waveform labeled 316occurs with a low bias that results in a short conduction time. Thiscondition provides minimum output. The waveform labeled (320) occurswith a high bias input and corresponds to a large conduction angle andprovides maximum output. A simultaneous optimization of all parametersis also possible.

[0170] Regulation on the Primary Side (with a Dual Switch):

[0171]FIGS. 3-8 and 3-12 show other arrangements to provide regulationon the primary side of the transformer. This circuit can use twoswitches (323) that may operate 180 degrees out of phase. They canoperate so as to move from a series resonance between a capacitor (321)and the leakage inductance (322) of the series transformer (320). Thisoccurs when both switches are closed. This shorts the primary inductanceand leaves only a series resonance already mentioned. This condition cangive maximum AC voltage to the rectifier circuit.

[0172] A second condition can occur when both switches are completelyopen. During this condition the capacitors (324) (which includes theswitch adjunct capacitance) can be in series across the series switchtransformer. It is also possible to just use a capacitor across thetransformer (325) or a combination of both. This total capacitance canbe resonant with the magnetizing inductance of the transformer. This cancreate a parallel resonant circuit in series with the primary of themain transformer and may result in minimum AC voltage to the rectifiercircuit.

[0173] The third and normal condition can occur with a variableconduction angle. With the values disclosed this circuit can operateover the entire conduction range with ZVS.

[0174] Natural Regulation:

[0175] If certain values of total inductance, total capacitance and theoutput filter inductance are chosen correctly a new phenomenon canexist. The DC output voltage can remain relatively independent of theload current. This can occur without any variable elements or feedback.

[0176] Examples:

[0177] Choosing all the circuit parametric values can be a lengthy task.The following example is a general-purpose rectifier which may beoptimized for powering a microprocessor operating at 1.8 volts andrequiring 20 amperes. Using the circuit of FIG. 3-3 the followingparametric values may be appropriate: Frequency = 3.3 MHZ Turns ratio =5:1 Input voltage = 30 VAC LT = 30 nH CT = 10 nF Cin = 2 nF L1 & L2 =100 nH Co = 500 μF SR 1 & SR 2 = 3 ea. FDS6880 Conduction angle = 266degrees Delay angle = 24 degrees

[0178]FIG. 3-5 shows one embodiment of a SR gate drive; it consists ofsummation of sinusoidal signal derived from the AC input plus a controlsignal. Also, the signal derived from the AC input can have an optimaldelay for high efficiency. This circuit can produce a clean AC voltageby taking advantage of the gate transformer leakage inductance and thegate capacitance to filter harmonics from the AC input. This circuit canalso show the creation of delay using R1,2, the combination of C1,2(which includes the adjunct gate to source capacitance), and theinductor L1.

[0179] Output Trap:

[0180] Also shown in FIG. 5 is a valuable filter element. C3 and L1 canform a parallel circuit resonant at twice the fundamental frequency.This parallel trap can provide the following advantages:

[0181] 1.1) targeting largest ripple component only

[0182] 2) storing very little energy—allowing fast loop control

[0183] 3) sharply reducing the AC current component of the connection tothe output capacitor.

[0184] If this circuit powers a microprocessor, the C4 may be criticallylocated to minimize inductance to the microprocessor. In this case theparallel trap can minimize the ‘hot leads’ problem for the connectionfrom the rest of the circuit to the C out.

[0185] Topology Variations:

[0186]FIGS. 3-9 A, B, C, and D show various topologies that may be usedto implement the invention disclosed. The location of the totalinductance and total capacitance is shown in each. FIG. 3-9 A shows asingle ended version. This can be an excellent topology for low costconcerns. FIG. 3-9 B shows the effect of a transformer with a centertap. This circuit can be useful but may not utilize the transformersecondary fully. In addition for low voltages some realizations canrequire the secondary to have only one turn possibly making a center tapmore difficult to implement. FIG. 3-9 C shows inverting the SR's and thefilter inductors. This circuit can be almost identical to the preferredone. In addition, the gate drive may not be referenced to a commonsource point making the drive circuit more complex (not shown). FIG. 3-9D shows a center tapped coil in place of a center tapped secondary. Somemagnetic realizations make this circuit attractive. The essentials ofthis disclosure apply as well.

[0187] The above examples represent only a few of the many designspossible. It should be obvious from these variations that other circuitsmay be designed which embody the ideas disclosed.

[0188] Third Harmonic Trap:

[0189] As may be understood from the above and the circuit designs, evenor odd harmonics may exist or be of concern in different directions. Forexamples even order harmonics (i.e. 2nd, 4th, etc.) may be of concern inthe forward direction and odd order harmonics (i.e. 3rd, 5th, etc.) maybe of concern in the backward direction. Each may be addressed.Naturally, the highest order of such harmonics (i.e. 2nd or 3rd) may beof initial interest. In the above discussion, a forward concerned, evenorder harmonic (e.g. the 2nd harmonic) was addressed. A backwardconcerned, odd order harmonic (e.g. the 3rd harmonic) may also beaddressed. For the third harmonic, a series connection of an inductorand capacitor tuned to the third harmonic can be placed across theprimary of the main VRM transformer. The preferred embodiment disclosedcan draw an input current with substantial third harmonic content. Byplacing a trap on the input of the circuit the harmonic currents canflow through the trap and may not appear on the distribution supplyingthe circuit. As those skilled in the art would easily appreciate, bysimple tuning, other harmonics can also be addressed.

[0190] More importantly, the efficiency of the rectifier can be improvedwith the addition of a third harmonic trap. The output circuit can benon-linear especially with the SR's having a long conduction angle (seeFIG. 3-4).

[0191] The DC output voltage from this circuit (FIGS. 3-4 & 3-10) can beequal to the integral of the voltage across the SR's (the averagevoltage across an inductor must be zero). Any distortion of thiswaveform can usually cause a reduction of the DC output voltage andconsequently a reduction in efficiency. The third harmonic trap canpreserve the natural peak of the SR voltage waveform.

[0192] Another potential benefit of the third harmonic trap is improvedstability of a system where multiple SR circuits are fed from a commonAC source. A local third harmonic trap can prevent SR circuits frominteracting due to third harmonic current flowing along the distributionpath.

[0193] Better put, without a third harmonic trap negative impedance canexist during a SR non-conduction time. Slight phase variations betweenSR circuits can result in high harmonic energy flowing between SRcircuits. This can manifest itself in overall system instability. Thepresence of a third harmonic trap on the input of each SR circuit canlocally satisfy the high order current requirement and can result insystem stability.

[0194] Remote Power:

[0195] Devices like microprocessors can require low voltage, highcurrent and exhibit high di/dt requirements. In the circuit of FIG.3-10, one problem which can exist is the di/dt limitation caused by theinterconnect inductance (326). In this commonly used circuit, bypasscapacitor (328) (which may be composed by many small capacitors inparallel) can be located near the microprocessor power pins. A largercapacitor, often called the bulk capacitor (327), can be located a smalldistance away. The short distance between capacitors (327) and (328) canform an inductor (326). This inductor (326) may limit the maximum di/dtthe microprocessor can pull from the power supply. This can beespecially true if the bypass capacitor is small (this is normally thecase) and/or the basic power conversion frequency is too low (also thenormal case). The bypass capacitor (328) may not be kept charged to thedemanded voltage. Even if the power supply feeding capacitor (327) wereideal, or if capacitor (327)were replaced with an ideal voltage source adi/dt limit might still exist as a result of the interconnect inductance(326).

[0196] In the circuit of the invention this problem can be overcome.Referring to FIG. 3-3, with this method and circuit the power conversionfrequency can be increased to the point where the output capacitance canbe small enough to be used as the microprocessor bypass capacitor whichcan be located adjacent to the microprocessor power pins; hence theoutput can be substantially non-capacitive. Thus the DC supply voltagefor the particular component can be located electrically remote to thecomponent itself. This location can avoid the need for providing the VRMimmediately adjacent to the particular component involved. Importantly,with the present invention, the DC voltage can now be supplied atdistances such as greater than about one-half an inch from the activeportion (such as the microprocessor itself) of the component. Byconsidering the active portion of the component, that is, the portionwhich consumes the power to achieve some desired function—other thanmerely transmitting the power such as wires or connectors or the likedo, the true electrical effect of being remotely located can be fullyappreciated. Significantly, with this design even greater distances forlocating the power are possible. This may include distances of not onlygreater than about one-half an inch from the active portion, but alsodistances of greater than about one inch from the active portion andeven distances of greater than about two inches from the active portion.

[0197] Quiet Power:

[0198] One of the problems facing the power supply industry as voltagesdrop, currents increase and di/dt requirements increase is noise. Thecircuit of FIG. 3-1 is noisy for three reasons.

[0199] First the switching FET's (329) may be force commutated withsteep voltage wavefronts. This can conduct and radiate noise into thesurrounding structures. Compare the voltage waveforms of FIG. 3-2 tothose of FIG. 3-4 to see the difference.

[0200] Second, the input circuit shown in FIG. 3-1 can inject currentinto the ground path. As the FET's (329) are switched, large current canflow around loop (330)through the input capacitor (332), interconnectinductance (331) and FET's (329). The rate of change of current di/dtaround this loop (330) can cause a voltage to be developed acrossinductor (331) which can be impressed onto the output voltage.

[0201] Third, the output of a circuit like FIG. 3-1 can be inherentlynoisy as the DC output voltage is reduced. The DC output voltage is theaverage value of the voltage on point 2 shown in FIG. 3-2. The voltageregulation method is sometimes dubbed pulse width modulation. For loweroutput voltages the pulse width becomes narrower to the point ofdifficulty of control. This is because a variation in width is a largerpercentage of the total pulse width. This can create a shaky or noisyoutput voltage.

[0202] The circuits being disclosed can use zero voltage switching (ZVS)and can have smooth voltage waveforms in the rectification circuitry.Compare the voltage waveforms for FIG. 3-2 (Prior Art) to FIG. 3-4. Itis obvious the waveforms on the invention will be less noisy. Secondlyin one preferred embodiment the regulation can occur on the primary ofthe transformer. This circuit is also ZVS plus it is isolated from theDC output voltage. These factors combined can make this approach muchmore suited to the next generation low voltage devices.

[0203] Additional Example:

[0204]FIGS. 3-11 and 3-12 show schematics for a complete AC to DC powerconverter which can include the rectifier section, the gate drive, theseries switch(es) along with a self derived DCpower supply and feedbackfrom the output to the series switch for regulation. These schematicscan embody much of what has been disclosed and can show a completeworking 1.8 volt, 20 ampere DCpower supply suitable for loads requiringhigh di/dt. They can operate from an AC input buss at 30 volts RMS at afrequency of 3.39 MHZ. Finally, FIG. 3-13 shows a potential design forsome significant overall portions of the “silver box” as it may beconfigured in one preferred design.

[0205] Regulation on the Primary Side (with a Variable Capacitor):

[0206] The difference between a series switch on the primary side of thetransformer and a capacitor is that the capacitor can present a losslesselement. It may also be a linear element. Referring to the embodimentshown in FIG. 3-14, the variable capacitor (C1) can create a phase shiftbetween the primary AC energy source and the primary winding of the maintransformer. In this configuration of the primary side regulator themechanism of regulation is different from the one described previouslyfor single and dual switches. No resonance of the magnetizing inductanceis involved for the process of regulation. The primary elements ofregulation for this topology may include the gate drive phase angle andthe combination of series capacitor impedance with SR input impedance.Certain combinations of values of the series capacitor, the leakageinductance of the transformer(s), and the natural or additionalcapacitances of the SRs can provide a number of advantages including:

[0207] 1) The circuit can be relatively insensitive to the magnetizationinductance of the transformer (e.g. the stability of magneticpermeability of materials used for transformers can be largelyirrelevant);

[0208] 2) The phase delay circuit for the gate drive of the SR may nolonger be required, (e.g. elements L1, R1, R2, C1 and C2 as shown inFIG. 3-5 can be excluded);

[0209] 3) In situations of a variable load, while undergoing thevariable load conditions (e.g. an output current change) the SR gatedrive voltage can adjust automatically to the most efficient value forthe given load condition. For example, in one of the practicalrealizations of this circuit the efficiency at 10% current load was only15% less than at full load!

[0210] 4) The reactive part of the circuit can become constant underdifferent load conditions and may be brought to zero (for seriesequivalent R-X circuit) by adding a parallel inductor to the input ofthe circuit. That is, the input impedance of the circuit can staysubstantially non-reactive for the full range of load conditions. Thisis shown in FIG. 3-15 for various load conditions. This aspect can beimportant for the primary energy source as most AC generators can workefficiently only into substantially resistive loads. This feature canallow the use of a less complicated AC generator for the primary powersource.

[0211] 5) the phenomenon of natural regulation can appear. This canresult in limiting the range required for the series capacitor toachieve the full range of load regulation. For example, in oneembodiment, the series capacitor value range needed is only ±25% of themean value. A simple varactor element may be used to achieve this.

[0212] Regulation on the Primary Side (with a Switch Equivalent of theVariable Capacitor):

[0213] As a result of the limited range of the capacitance required aZVS switch can be used as an analog equivalent on the primary side oftransformer. The configuration of one realization of the switchequivalent can be similar to that described above with regard to FIG.3-8, but it operates in a different mode. This circuit can use twoswitches that may operate 180 degrees out of phase as can be understoodfrom FIG. 3-12. The circuit may be galvanically isolated from the SRwith a transformer. There may be no special requirements for thetransformer except in many cases it may need to have stable leakageinductance. The leakage inductance value can also be taken into accountduring circuit design and compensated if necessary. Neithermagnetization inductance nor leakage inductance may need to be part of aresonant circuit. There also may be no special requirements forstability of the core magnetic permeability for the transformer. Withproperly chosen circuit parameters ZVS switching and equivalence to thelinear variable capacitor can also be maintained over the wholeregulation range. Control of the value of effective capacitance may beset by the control DC bias voltage on the FET gates. In contrast to theSeries Switch embodiment described above, the waveform across theinsulation transformer may be substantially sinusoidal over the wholeregulation range and the amplitude may only change under a differentload condition.

[0214] Output Transformer:

[0215] Yet another potentially independent aspect of the invention isshown in FIG. 3-14. This shows another option for the output filterelement for the SR. Instead of two output inductors such as Lf shown inFIG. 3-3, only one transformer with 1:1 ratio can be used. Moregenerally, the output transformer may simply be two output inductances(W3 and W4 in FIG. 3-14) which are coupled in some manner. By using amagnetic coupling or even a transformer, the following advantages can berealized:

[0216] 1) Only one magnetic element instead of two may be used;

[0217] 2) The fundamental frequency AC current through the magneticelements may be sharply reduced, reducing also the radiated AC magneticfield;

[0218] 3) Leakage inductance of the transformer may b used as afiltering element for the output of the SR. Again, leakage inductance inthe first approach may not depend on magnetic permeability of the corehence no special requirements for magnetic material stability;

[0219] 4) The output DC current from the two halves of the SR may flowthrough the transformer in opposite directions and cancel each other sothe resulting DC magnetic field in the transformer core may be nearlyzero. As a result there may be no magnetic saturation in the core and asmall amount of magnetic material can be used in a closed configuration(toroid).

[0220] The discussion included in this patent is intended to serve as abasic description. The reader should be aware that the specificdiscussion may not explicitly describe all embodiments possible; manyalternatives are implicit. It also may not fully explain the genericnature of the invention and may not explicitly show how each feature orelement can actually be representative of a broader function or of agreat variety of alternative or equivalent elements. A variety ofchanges may be made without departing from the essence of the invention.All these are implicitly included in this disclosure. Where theinvention is described in device-oriented terminology, each element ofthe device implicitly performs a function. Apparatus claims are includedfor many of the embodiments described, however only initial methodclaims are presented. Both additional method claims to track theapparatus claims presented and even additional method and/or apparatusto address the various functions the invention and each element performsmay be included. Product by process claims or the like may also be addedto any results achieved through such systems. Importantly, it should beunderstood that neither the description, nor the terminology, nor thespecific claims presented is intended to limit the scope of the patentdisclosure or the coverage ultimately available. Coverage for computersystem as well as other electronics items may be presented and should beunderstood as encompassed by this application regardless of what isinitially presented or the title indicated. All this should beparticularly noted with respect to the method claims as well. Althoughclaims directed to the apparatus have been included in various detail,for administrative efficiencies, only initial claims directed toward themethods have been included. Naturally, the disclosure and claiming ofthe apparatus focus in detail is to be understood as sufficient tosupport the full scope of both method and apparatus claims. Additionalmethod claims may and likely will be added at a later date whenappropriate to explicitly claim such details. Thus, the presentdisclosure is to be construed as encompassing the full scope of methodclaims, including but not limited to claims and subclaims similar tothose presented in a apparatus context. In addition other claims forembodiments disclosed but not yet claimed may be added as well.

[0221] Further, the use of the principles described herein may result ina wide variety of configurations and, as mentioned, may permit a widevariety of design tradeoffs. In addition, each of the various elementsof the invention and claims may also be achieved in a variety of mannersor may be presented independently. This disclosure should be understoodto encompass each such variation and the various combinations andpermutations of any and all elements or applications. Particularly, itshould be understood that as the disclosure relates to elements of theinvention, the words for each element may be expressed by equivalentapparatus terms or method terms—even if only the function or result isthe same. Such equivalent, broader, or even more generic terms should beconsidered to be encompassed in the description of each element oraction. Such terms can be substituted where desired to make explicit theimplicitly broad coverage to which this invention is entitled. As butone example, it should be understood that all action may be expressed asa means for taking that action or as an element which causes thataction. Similarly, each physical element disclosed should be understoodto encompass a disclosure of the action which that physical elementfacilitates. Regarding this last aspect, the disclosure of a “switch”should be understood to encompass disclosure of the act of“switching”—whether explicitly discussed or not—and, conversely, werethere only disclosure of the act of “switching”, such a disclosureshould be understood to encompass disclosure of a “switch” or even a“means for switching.” Such changes and alternative terms are to beunderstood to be explicitly included in the description as isparticularly true for the present invention since its basic concepts andunderstandings are fundamental in nature and can be applied in a varietyof ways to a variety of fields.

[0222] Furthermore, any references mentioned in the application for thispatent as well as all references listed in any list of references filedwith the application are hereby incorporated by reference, however, tothe extent statements might be considered inconsistent with thepatenting of this invention such statements are expressly not to beconsidered as made by the applicant.

[0223] Finally, unless the context requires otherwise, the word“comprise” or variations such as “comprises” or “comprising”, should beunderstood to imply the inclusion of a stated element or step or groupof elements or steps but not the exclusion of any other element or stepor group of elements or steps. Additionally, the various combinationsand permutations of all elements or applications can be created andpresented. All can be done to optimize performance in a specificapplication.

1.-378. (canceled).
 379. A power supply circuit powering a low voltage,high current microprocessor capable of a rapid current demand comprisinga DC power supply having a substantially inductive DC output.
 380. Apower supply circuit powering a microprocessor as described in claim 379wherein said microprocessor comprises a low voltage, high current loadand wherein said DC power supply provides a regulated voltage to saidload.
 381. A power supply circuit powering a microprocessor as describedin claim 379 wherein said DC power supply is physically remote from saidmicroprocessor.
 382. A power supply circuit powering a microprocessor asdescribed in claim 381 wherein said DC power supply provides saidmicroprocessor power remotely over a distance selected from a groupconsisting of over at least about one-half inch from said DC powersupply to said microprocessor, over at least about one inch from said DCpower supply to said microprocessor, and over at least about two inchesfrom said DC power supply to said microprocessor.
 383. A power supplycircuit powering a microprocessor as described in claim 379 wherein saidDC power supply is electrically remote from said microprocessor.
 384. Apower supply circuit powering a microprocessor as described in claim 379further comprising a bypass capacitance adjacent said microprocessor.385. A power supply circuit powering a microprocessor as described inclaim 384 wherein said bypass capacitance comprises a total bypasscapacitance selected from a group consisting of less than about 0.2millifarads and less than about 0.5 millifarads.
 386. A power supplycircuit powering a microprocessor as described in claim 384 wherein saidbypass capacitance comprises a capacitance selected from a groupconsisting of less than about 0.3 millifarads, less than about 0.5millifarads, less than about 1 millifarads, less than about 3millifarads, less than about 10 millifarads, about only the inherentcapacitance of a response network, about only an inherent reactance of acomponent connector, about only an inherent capacitance of a lowvoltage, high current component, about only a bypass capacitance of amicroprocessor, and any permutations or combinations of the above. 387.A power supply circuit powering a microprocessor as described in claim379 wherein said substantially inductive DC output comprises asubstantially non-capacitive output.
 388. A power supply circuitpowering a microprocessor as described in claim 380 wherein saidmicroprocessor comprises a microprocessor operating at a nominal DCvoltage selected from a group consisting of less than about 2 volts,less than about 1.8 volts, less than about 1.5 volts, less than about1.3 volts, less than about 1 volt, and less than about 0.4 volts.
 389. Apower supply circuit powering a microprocessor as described in claim 380wherein said microprocessor is capable of a rapid current demand whichrises at a level selected from a group consisting of at least about 0.2amperes per nanosecond, at least about 0.5 amperes per nanosecond, atleast about 1 ampere per nanosecond, at least about 3 amperes pernanosecond, at least about 10 amperes per nanosecond, and at least about30 amperes per nanosecond.
 390. A power supply circuit powering amicroprocessor as described in claim 380 wherein said microprocessorcomprises a microprocessor operating at a maximum current selected froma group consisting of more than about 15 amperes, more than about 20amperes, more than about 50 amperes, and more than about 100 amperes.391. A power supply circuit powering a microprocessor as described inclaim 379 wherein said DC power supply comprises a voltage regulationmodule.
 392. A method of powering a low voltage, high currentmicroprocessor capable of a rapid current demand, comprising the stepsof: a. providing a DC power supply having a substantially inductive DCoutput; and b. powering said microprocessor with said substantiallyinductive DC output.
 393. A method of powering a low voltage, highcurrent microprocessor as described in claim 392 wherein said step ofpowering said microprocessor with said substantially inductive DC outputcomprises the step of powering a low voltage, high currentmicroprocessor.
 394. A method of powering a low voltage, high currentmicroprocessor as described in claim 392 wherein said step of providinga DC power supply having a substantially inductive DC output comprisesthe step of providing a DC power supply physically remote from saidmicroprocessor.
 395. A method of powering a low voltage, high currentmicroprocessor as described in claim 394 wherein said step powering saidmicroprocessor with said substantially inductive DC output comprises thestep of transmitting said substantially inductive DC output over adistance selected from a group consisting of over at least aboutone-half inch, over at least about one inch, and over at least about twoinches.
 396. A method of powering a low voltage, high currentmicroprocessor as described in claim 392 wherein said step of poweringsaid microprocessor with said substantially inductive DC outputcomprises the step of transmitting said substantially inductive DCoutput to an electrically remote location.
 397. A method of powering alow voltage, high current microprocessor as described in claim 392further comprising the step of establishing a bypass capacitanceadjacent said microprocessor.
 398. A method of powering a low voltage,high current microprocessor as described in claim 397 wherein said stepof establishing a bypass capacitance adjacent said microprocessorcomprises the step of establishing a total bypass capacitance selectedfrom a group consisting of less than about 0.2 millifarads and less thanabout 0.5 millifarads.
 399. A method of powering a low voltage, highcurrent microprocessor as described in claim 397 wherein said step ofpowering said microprocessor comprises the step of transmitting saidsubstantially inductive DC output through a substantially non-capacitiveDC output system having an effective capacitance selected from a groupconsisting of less than about 0.3 millifarads, less than about 0.5millifarads, less than about 1 millifarads, less than about 3millifarads, less than about 10 millifarads, about only the inherentcapacitance of a response network, about only an inherent reactance of acomponent connector, about only an inherent capacitance of said computercomponent, about only a bypass capacitance of a microprocessor, and anypermutations or combinations of the above.
 400. A method of powering alow voltage, high current microprocessor as described in claim 392wherein said step of powering said microprocessor with saidsubstantially inductive DC output comprises the step of powering saidcomputer component from said component DC supply voltage through asubstantially non-capacitive DC output system.
 401. A method of poweringa low voltage, high current microprocessor as described in claim 393wherein said step of powering said microprocessor with saidsubstantially inductive DC output comprises the step of transmittingsaid substantially inductive DC output at a nominal DC voltage selectedfrom a group consisting of less than about 2 volts, less than about 1.8volts, less than about 1.5 volts, less than about 1.3 volts, less thanabout 1 volt, and less than about 0.4 volts.
 402. A method of powering alow voltage, high current microprocessor as described in claim 393wherein said step of powering said microprocessor with saidsubstantially inductive DC output comprises the step of transmittingsaid substantially inductive DC output through a DC output systemcapable of a rapid current demand which rises at a level selected from agroup consisting of at least about 0.2 amperes per nanosecond, at leastabout 0.5 amperes per nanosecond, at least about 1 ampere pernanosecond, at least about 3 amperes per nanosecond, at least about 10amperes per nanosecond, and at least about 30 amperes per nanosecond.403. A method of powering a low voltage, high current microprocessor asdescribed in claim 393 wherein said step of powering said microprocessorwith said substantially inductive DC output comprises the step oftransmitting said substantially inductive DC output through a DC outputsystem operating at a maximum current selected from a group consistingof more than about 15 amperes, more than about 20 amperes, more thanabout 50 amperes, and more than about 100 amperes.
 404. A method ofpowering a low voltage, high current microprocessor as described inclaim 392 wherein said step of providing a DC power supply having asubstantially inductive DC output comprises the step of providing a DCpower supply that comprises a voltage regulation module.
 405. A powersupply circuit powering a low voltage, high current microprocessorcapable of a rapid current demand comprising a voltage regulation modulehaving a substantially non-capacitive DC output.
 406. A power supplycircuit powering a microprocessor as described in claim 405 wherein saidmicroprocessor comprises a low voltage, high current load and whereinsaid voltage regulation module provides a regulated voltage to saidload.
 407. A power supply circuit powering a microprocessor as describedin claim 405 wherein said voltage regulation module is physically remotefrom said microprocessor.
 408. A power supply circuit powering amicroprocessor as described in claim 407 wherein said voltage regulationmodule provides said microprocessor power remotely over a distanceselected from a group consisting of over at least about one-half inchfrom said voltage regulation module to said microprocessor, over atleast about one inch from said voltage regulation module to saidmicroprocessor, and over at least about two inches from said voltageregulation module to said microprocessor.
 409. A power supply circuitpowering a microprocessor as described in claim 405 wherein said voltageregulation module is electrically remote from said microprocessor. 410.A power supply circuit powering a microprocessor as described in claim405 further comprising a bypass capacitance adjacent saidmicroprocessor.
 411. A power supply circuit powering a microprocessor asdescribed in claim 410 wherein said bypass capacitance comprises a totalbypass capacitance selected from a group consisting of less than about0.2 millifarads and less than about 0.5 millifarads.
 412. A power supplycircuit powering a microprocessor as described in claim 410 wherein saidbypass capacitance comprises a capacitance selected from a groupconsisting of less than about 0.3 millifarads, less than about 0.5millifarads, less than about 1 millifarads, less than about 3millifarads, less than about 10 millifarads, about only the inherentcapacitance of a response network, about only an inherent reactance of acomponent connector, about only an inherent capacitance of a lowvoltage, high current component, about only a bypass capacitance of amicroprocessor, and any permutations or combinations of the above. 413.A power supply circuit powering a microprocessor as described in claim405 wherein said substantially non-capacitive DC output comprises asubstantially inductive DC output.
 414. A power supply circuit poweringa microprocessor as described in claim 413 wherein a substantiallyinductive DC output comprises an inductance internal to said voltageregulation module.
 415. A power supply circuit powering a microprocessoras described in claim 414 wherein said inductance internal to saidvoltage regulation module comprises an inductance selected from a groupconsisting of a total series inductance and an interconnect inductance.416. A power supply circuit powering a microprocessor as described inclaim 413 or 414 wherein said substantially inductive DC outputcomprises an inductance external to said voltage regulation module. 417.A power supply circuit powering a microprocessor as described in claim406 wherein said microprocessor comprises a microprocessor operating ata nominal DC voltage selected from a group consisting of less than about2 volts, less than about 1.8 volts, less than about 1.5 volts, less thanabout 1.3 volts, less than about 1 volt, and less than about 0.4 volts.418. A power supply circuit powering a microprocessor as described inclaim 406 wherein said microprocessor is capable of a rapid currentdemand which rises at a level selected from a group consisting of atleast about 0.2 amperes per nanosecond, at least about 0.5 amperes pernanosecond, at least about 1 ampere per nanosecond, at least about 3amperes per nanosecond, at least about 10 amperes per nanosecond, and atleast about 30 amperes per nanosecond.
 419. A power supply circuitpowering a microprocessor as described in claim 406 wherein saidmicroprocessor comprises a microprocessor operating at a maximum currentselected from a group consisting of more than about 15 amperes, morethan about 20 amperes, more than about 50 amperes, and more than about100 amperes.
 420. A method of powering a low voltage, high currentmicroprocessor capable of a rapid current demand, comprising the stepsof: a. providing a voltage regulation module having a substantiallynon-capacitive DC output; and b. powering said microprocessor with saidsubstantially non-capacitive DC output.
 421. A method of powering a lowvoltage, high current microprocessor as described in claim 420 furthercomprising the step of regulating a voltage of said substantiallynon-capacitive DC output with said voltage regulation module, andwherein said step of powering said microprocessor with saidsubstantially non-capacitive DC output comprises the step of poweringsaid microprocessor with a low voltage, high current load.
 422. A methodof powering a low voltage, high current microprocessor as described inclaim 420 further comprising the step of establishing said voltageregulation module physically remote from said microprocessor; andwherein said step of powering said microprocessor with saidsubstantially non-capacitive DC output comprises the step of remotelypowering said microprocessor.
 423. A method of powering a low voltage,high current microprocessor as described in claim 422 wherein said stepof remotely powering said microprocessor comprises the step oftransmitting said substantially non-capacitive DC output over a distanceselected from a group consisting of over at least about one-half inchfrom said voltage regulation module to said microprocessor, over atleast about one inch from said voltage regulation module to saidmicroprocessor, and over at least about two inches from said voltageregulation module to said microprocessor.
 424. A method of powering alow voltage, high current microprocessor as described in claim 420wherein said step of powering said microprocessor with saidsubstantially non-capacitive DC output comprises the step oftransmitting said substantially non-capacitive DC output to anelectrically remote microprocessor
 425. A method of powering a lowvoltage, high current microprocessor as described in claim 420 furthercomprising the step of establishing a bypass capacitance adjacent saidmicroprocessor.
 426. A method of powering a low voltage, high currentmicroprocessor as described in claim 425 wherein said step ofestablishing a bypass capacitance adjacent said microprocessor comprisesthe step of establishing a total bypass capacitance selected from agroup consisting of less than about 0.2 millifarads and less than about0.5 millifarads.
 427. A method of powering a low voltage, high currentmicroprocessor as described in claim 425 wherein said step oftransmitting said substantially non-capacitive DC output comprises thestep of transmitting said substantially non-capacitive DC output througha substantially non-capacitive DC output system having an effectivecapacitance selected from a group consisting of less than about 0.3millifarads, less than about 0.5 millifarads, less than about 1millifarads, less than about 3 millifarads, less than about 10millifarads, about only the inherent capacitance of a response network,about only an inherent reactance of a component connector, about only aninherent capacitance of said computer component, about only a bypasscapacitance of a microprocessor, and any permutations or combinations ofthe above.
 428. A method of powering a low voltage, high currentmicroprocessor as described in claim 420 wherein said step of providinga voltage regulation module having a substantially non-capacitive DCoutput comprises the step of providing a voltage regulation modulehaving a substantially inductive DC output.
 429. A method of powering alow voltage, high current microprocessor as described in claim 428wherein said step of providing a voltage regulation module having asubstantially inductive DC output comprises the step of providing avoltage regulation module having an inductance internal to said voltageregulation module.
 430. A method of powering a low voltage, high currentmicroprocessor as described in claim 429 wherein said step of providinga voltage regulation module having an inductance internal to saidvoltage regulation module comprises the step of providing a voltageregulation module having an inductance selected from a group consistingof a total series inductance and an interconnect inductance.
 431. Amethod of powering a low voltage, high current microprocessor asdescribed in claim 425 or 426 wherein said step of providing a voltageregulation module having a substantially non-capacitive DC outputcomprises the step of providing a voltage regulation module having aninductance external to said voltage regulation module.
 432. A method ofpowering a low voltage, high current microprocessor as described inclaim 421 wherein said step of powering said microprocessor comprisesthe step of powering said microprocessor from said component DC supplyvoltage at a nominal DC voltage selected from a group consisting of lessthan about 2 volts, less than about 1.8 volts, less than about 1.5volts, less than about 1.3 volts, less than about 1 volt, and less thanabout 0.4 volts.
 433. A method of powering a low voltage, high currentmicroprocessor as described in claim 421 wherein said step of poweringsaid microprocessor comprises the step of transmitting a low voltage,high current load through a DC output system capable of a rapid currentdemand which rises at a level selected from a group consisting of atleast about 0.2 amperes per nanosecond, at least about 0.5 amperes pernanosecond, at least about 1 ampere per nanosecond, at least about 3amperes per nanosecond, at least about 10 amperes per nanosecond, and atleast about 30 amperes per nanosecond.
 434. A method of powering a lowvoltage, high current microprocessor as described in claim 421 whereinsaid step of powering said microprocessor with a low voltage, highcurrent load comprises the step of transmitting said low voltage, highcurrent load through a DC output system operating at a maximum currentselected from a group consisting of more than about 15 amperes, morethan about 20 amperes, more than about 50 amperes, and more than about100 amperes.
 435. A power supply circuit powering a low voltage, highcurrent microprocessor capable of a rapid current demand comprising avoltage regulation module having a substantially inductive DC output.436. A power supply circuit powering a microprocessor as described inclaim 435 wherein said microprocessor comprises a low voltage, highcurrent load and wherein said voltage regulation module provides aregulated voltage to said load.
 437. A power supply circuit powering amicroprocessor as described in claim 435 wherein said voltage regulationmodule is physically remote from said microprocessor.
 438. A powersupply circuit powering a microprocessor as described in claim 437wherein said voltage regulation module provides said microprocessorpower remotely over a distance selected from a group consisting of overat least about one-half inch from said voltage regulation module to saidmicroprocessor, over at least about one inch from said voltageregulation module to said microprocessor, and over at least about twoinches from said voltage regulation module to said microprocessor. 439.A power supply circuit powering a microprocessor as described in claim435 wherein said voltage regulation module is electrically remote fromsaid microprocessor.
 440. A power supply circuit powering amicroprocessor as described in claim 435 further comprising a bypasscapacitance adjacent said microprocessor.
 441. A power supply circuitpowering a microprocessor as described in claim 440 wherein said bypasscapacitance comprises a total bypass capacitance selected from a groupconsisting of less than about 0.2 millifarads and less than about 0.5millifarads.
 442. A power supply circuit powering a microprocessor asdescribed in claim 440 wherein said bypass capacitance comprises acapacitance selected from a group consisting of less than about 0.3millifarads, less than about 0.5 millifarads, less than about 1millifarads, less than about 3 millifarads, less than about 10millifarads, about only the inherent capacitance of a response network,about only an inherent reactance of a component connector, about only aninherent capacitance of a low voltage, high current component, aboutonly a bypass capacitance of a microprocessor, and any permutations orcombinations of the above.
 443. A power supply circuit powering amicroprocessor as described in claim 435 wherein said substantiallyinductive DC output comprises a substantially non-capacitive DC output.444. A power supply circuit powering a microprocessor as described inclaims 435 wherein said substantially inductive DC output comprises aninductance internal to said voltage regulation module.
 445. A powersupply circuit powering a microprocessor as described in claim 444wherein said inductance internal to said voltage regulation modulecomprises an inductance selected from a group consisting of a totalseries inductance and an interconnect inductance.
 446. A power supplycircuit powering a microprocessor as described in claims 435 or 423wherein said substantially inductive DC output comprises an inductanceexternal to said voltage regulation module.
 447. A power supply circuitpowering a microprocessor as described in claim 436 wherein saidmicroprocessor comprises a microprocessor operating at a nominal DCvoltage selected from a group consisting of less than about 2 volts,less than about 1.8 volts, less than about 1.5 volts, less than about1.3 volts, less than about 1 volt, and less than about 0.4 volts.
 448. Apower supply circuit powering a microprocessor as described in claim 436wherein said microprocessor is capable of a rapid current demand whichrises at a level selected from a group consisting of at least about 0.2amperes per nanosecond, at least about 0.5 amperes per nanosecond, atleast about 1 ampere per nanosecond, at least about 3 amperes pernanosecond, at least about 10 amperes per nanosecond, and at least about30 amperes per nanosecond.
 449. A power supply circuit powering amicroprocessor as described in claim 436 wherein said microprocessorcomprises a microprocessor operating at a maximum current selected froma group consisting of more than about 15 amperes, more than about 20amperes, more than about 50 amperes, and more than about 100 amperes.450. A method of powering a low voltage, high current microprocessorcapable of a rapid current demand, comprising the steps of: a. providinga voltage regulation module having a substantially inductive DC output;and b. powering said microprocessor with said substantially inductive DCoutput.
 451. A method of powering a low voltage, high currentmicroprocessor as described in claim 450 further comprising the step ofregulating a voltage of said substantially inductive DC output with saidvoltage regulation module, and wherein said step of powering saidmicroprocessor with said substantially inductive DC output comprises thestep of powering said microprocessor with a low voltage, high currentload.
 452. A method of powering a low voltage, high currentmicroprocessor as described in claim 450 further comprising the step ofestablishing said voltage regulation module physically remote from saidmicroprocessor; and wherein said step of powering said microprocessorwith said substantially inductive DC output comprises the step ofremotely powering said microprocessor.
 453. A method of powering a lowvoltage, high current microprocessor as described in claim 452 whereinsaid step of remotely powering said microprocessor comprises the step oftransmitting said substantially inductive DC output over a distanceselected from a group consisting of over at least about one-half inchfrom said voltage regulation module to said microprocessor, over atleast about one inch from said voltage regulation module to saidmicroprocessor, and over at least about two inches from said voltageregulation module to said microprocessor.
 454. A method of powering alow voltage, high current microprocessor as described in claim 450wherein said step of powering said microprocessor with saidsubstantially inductive DC output comprises the step of transmittingsaid substantially inductive DC output to an electrically remotemicroprocessor.
 455. A method of powering a low voltage, high currentmicroprocessor as described in claim 450 further comprising the step ofestablishing a bypass capacitance adjacent said microprocessor.
 456. Amethod of powering a low voltage, high current microprocessor asdescribed in claim 455 wherein said step of establishing a bypasscapacitance adjacent said microprocessor comprises the step ofestablishing a total bypass capacitance selected from a group consistingof less than about 0.2 millifarads and less than about 0.5 millifarads.457. A method of powering a low voltage, high current microprocessor asdescribed in claim 455 wherein said step of establishing a bypasscapacitance adjacent said microprocessor comprises the step ofestablishing an effective capacitance selected from a group consistingof less than about 0.3 millifarads, less than about 0.5 millifarads,less than about 1 millifarads, less than about 3 millifarads, less thanabout 10 millifarads, about only the inherent capacitance of a responsenetwork, about only an inherent reactance of a component connector,about only an inherent capacitance of said computer component, aboutonly a bypass capacitance of a microprocessor, and any permutations orcombinations of the above.
 458. A method of powering a low voltage, highcurrent microprocessor as described in claim 450 wherein said step ofproviding a voltage regulation module having a substantially inductiveDC output comprises the step of providing a voltage regulation modulehaving a substantially non-capacitive DC output.
 459. A method ofpowering a low voltage, high current microprocessor as described inclaim 450 wherein said step of providing a voltage regulation modulehaving a substantially inductive DC output comprises the step ofproviding a voltage regulation module having an inductance internal tosaid voltage regulation module.
 460. A method of powering a low voltage,high current microprocessor as described in claim 459 wherein said stepof providing a voltage regulation module having an inductance internalto said voltage regulation module comprises the step of providing avoltage regulation module having an inductance selected from a groupconsisting of a total series inductance and an interconnect inductance.461. A method of powering a low voltage, high current microprocessor asdescribed in claim 450 or 459 wherein said step of providing a voltageregulation module having a substantially inductive DC output comprisesthe step of providing a voltage regulation module having an inductanceexternal to said voltage regulation module.
 462. A method of powering alow voltage, high current microprocessor as described in claim 451wherein said step of powering said microprocessor comprises the step ofpowering said microprocessor from said component DC supply voltage at anominal DC voltage selected from a group consisting of less than about 2volts, less than about 1.8 volts, less than about 1.5 volts, less thanabout 1.3 volts, less than about 1 volt, and less than about 0.4 volts.463. A method of powering a low voltage, high current microprocessor asdescribed in claim 451 wherein said step of powering said microprocessorcomprises the step of transmitting a low voltage, high current loadthrough a DC output system capable of a rapid current demand which risesat a level selected from a group consisting of at least about 0.2amperes per nanosecond, at least about 0.5 amperes per nanosecond, atleast about 1 ampere per nanosecond, at least about 3 amperes pernanosecond, at least about 10 amperes per nanosecond, and at least about30 amperes per nanosecond.
 464. A method of powering a low voltage, highcurrent microprocessor as described in claim 451 wherein said step ofpowering said microprocessor with a low voltage, high current loadcomprises the step of transmitting said low voltage, high current loadthrough a DC output system operating at a maximum current selected froma group consisting of more than about 15 amperes, more than about 20amperes, more than about 50 amperes, and more than about 100 amperes.